Lighting circuit of automotive lamp

ABSTRACT

A lighting circuit turns on a plurality of semiconductor light sources. Multiple current sources are each coupled to a corresponding semiconductor light source. A switching converter supplies a driving voltage V OUT  across each of multiple series connection circuits each formed of the semiconductor light source and the current source. A converter controller employing a ripple control method turns on a switching transistor of the switching converter in response to a voltage across any one of the multiple current sources decreasing to a bottom limit voltage.

BACKGROUND OF THE INVENTION 1. Field of the Invention

The present invention relates to a lighting circuit.

2. Description of the Related Art

Typical automotive lamps are capable of switching between a low-beammode and a high-beam mode. The low-beam mode is used to illuminate aclose range in the vicinity of the user's vehicle with a predeterminedlight intensity. In the low-beam mode, light distribution is determinedso as to prevent glare being imparted to an oncoming vehicle or aleading vehicle. The low-beam mode is mainly used when the vehicle istraveling in an urban area. In contrast, the high-beam mode is used toilluminate a distant range over a wide area ahead of the vehicle with arelatively high light intensity. The high-beam mode is mainly used whenthe vehicle is traveling at high speed along a road where there are asmall number of oncoming vehicles and leading vehicles. Accordingly, thehigh-beam mode provides the driver with high visibility, which is anadvantage, as compared with the low-beam mode. However, the high-beammode has a problem of imparting glare to a pedestrian or a driver of avehicle ahead of the vehicle.

In recent years, the ADB (Adaptive Driving Beam) technique has beenproposed in which a high-beam distribution pattern is dynamically andadaptively controlled based on the state of the surroundings of avehicle. With the ADB technique, the presence or absence of a leadingvehicle, an oncoming vehicle, or a pedestrian ahead of the vehicle isdetected, and the illumination is reduced or turned off for a regionthat corresponds to such a vehicle or pedestrian thus detected, therebyreducing glare imparted to such a vehicle or pedestrian.

FIG. 1 is a block diagram showing a lamp system 1001 having an ADBfunction. The lamp system 1001 includes a battery 1002, a switch 1004, aswitching converter 1006, multiple light-emitting units 1008_1 through1008_N, multiple current sources 1010_1 through 1010_N, a convertercontroller 1012, and a light distribution controller 1014.

The multiple light-emitting units 1008_1 through 1008_N are eachconfigured as a semiconductor light source such as an LED(light-emitting diode), LD (laser diode), or the like, which areassociated with multiple different regions on a virtual vertical screenahead of the vehicle. The multiple current sources 1010_1 through 1010_Nare arranged in series with the multiple corresponding light-emittingunits 1008_1 through 1008_N. A driving current I_(LED1) generated by thecurrent source 1010_i flows through the i-th (1 i≤N) light-emitting unit1008_i.

The multiple current sources 1010_1 through 1010_N are each configuredto be capable of turning on and off (or adjusting the amount of current)independently. The light distribution controller 1014 controls theon/off state (or the amount of current) for each of the multiple currentsources 1010_1 through 1010_N so as to provide a desired lightdistribution pattern.

The switching converter 1006 configured to provide a constant voltageoutput generates a driving voltage V_(OUT) that is sufficient for themultiple light-emitting units 1008_1 through 1008_N to provide lightemission with a desired luminance. Description will be made directingattention to the i-th channel. When a given driving current I_(LEDi)flows through the light-emitting unit 1008_i, a voltage drop (forwardvoltage) V_(Fi) occurs in the light-emitting unit 1008_i. In order toallow the current source 1010_i to generate the driving currentI_(LEDi), the voltage across the current source 1010_i is required to belarger than a particular voltage (which will be referred to as“V_(SATi)” hereafter). Accordingly, the following inequality expressionmust hold true.V _(OUT) >V _(Fi) +V _(SATi)  (1)

This relation must hold true for all the channels.

In order to satisfy the inequality expression (1) in all situations, theoutput voltage V_(OUT) may preferably be employed as the control targetfor the feedback control. Specifically, as represented by Expression(2), a target value V_(OUT(REF)) of the output voltage V_(OUT) is set toa higher value using a margin. Furthermore, the output voltage V_(OUT)may preferably be feedback controlled such that the output voltageV_(OUT) of the switching converter 1006 matches the target valueV_(OUT(REF)).V _(OUT(REF)) =V _(F(MARGIN)) +V _(SAT(MARGIN))

Here, V_(F(TYP)) represents the maximum value (or typical value) ofV_(F) with a margin added. V_(SAT(MARGIN)) represents a saturationvoltage V_(SAT) to which a margin is added.

In this control operation, the difference between the saturation voltageV_(SAT(MARGIN)) and the actual saturation voltage V_(SAT) is applied tothe current source 1010, which leads to the occurrence of unnecessarypower loss. In addition, when the actual forward voltage V_(F) is lowerthan V_(F(MARGIN)), voltage drop that occurs across the current source1010 includes the voltage difference between them, leading to theoccurrence of unnecessary power loss.

With an automotive lamp, there is a need to flow a very large currentthrough a light-emitting unit. Furthermore, it is more difficult toprovide such an automotive lamp with countermeasures for releasing heatthan it is for other devices. Accordingly, with the automotive lamp,there is a demand to reduce the heat amount due to the current source asmuch as possible.

SUMMARY OF THE INVENTION

The present invention has been made in order to solve such a problem.Accordingly, it is an exemplary purpose of an embodiment of the presentinvention to provide a lighting circuit that is capable of providingreduced power consumption.

A summary of several example embodiments of the disclosure follows. Thissummary is provided for the convenience of the reader to provide a basicunderstanding of such embodiments and does not wholly define the breadthof the disclosure. This summary is not an extensive overview of allcontemplated embodiments, and is intended to neither identify key orcritical elements of all embodiments nor to delineate the scope of anyor all aspects. Its sole purpose is to present some concepts of one ormore embodiments in a simplified form as a prelude to the more detaileddescription that is presented later.

1. An embodiment of the present invention relates to a lighting circuitstructured to turn on multiple semiconductor light sources. The lightingcircuit includes: multiple current sources each of which is to becoupled to a corresponding semiconductor light source; a switchingconverter structured to supply a driving voltage across each of multipleseries connection circuits each formed of the semiconductor light sourceand the current source; and a converter controller employing a ripplecontrol method. The converter controller turns on the switchingtransistor of the switching converter in response to a voltage acrossany one of the multiple current sources decreasing to a bottom limitvoltage.

2. An embodiment of the present invention also relates to the lightingcircuit structured to turn on multiple semiconductor light sources. Thelighting circuit includes: multiple current sources each of which is tobe coupled to a corresponding semiconductor light source in series, andeach of which includes a series transistor and a sensing resistorarranged in series with the corresponding semiconductor light source,and an error amplifier structured to adjust the voltage at the controlelectrode of the series transistor based on a voltage drop that occursacross the sensing resistor; a switching converter structured to supplya driving voltage across each of multiple series connection circuitseach formed of the semiconductor light source and the current source;and a converter controller using a ripple control method. The convertercontroller turns on a switching transistor of the switching converter inresponse to the output voltage of the error amplifier of any one of themultiple current sources satisfying a predetermined turn-on condition.

Another embodiment of the present invention relates to a current drivercircuit structured to drive multiple semiconductor light sources. Thecurrent driver circuit includes: multiple current sources eachstructured to allow the on/off state thereof to be controlledindependently according to a PWM signal, and to be each coupled to acorresponding semiconductor light source in series; an interface circuitstructured to receive, from an external processor at a first timeinterval, multiple control data that indicate an on/off duty cycle forthe multiple current sources; and a dimming pulse generator structuredto generate multiple PWM signals for the multiple current sources, andto gradually change, at a second time interval that is smaller than thefirst time interval, a duty cycle of each of the multiple PWM signalsfrom a value indicated by the corresponding control data before updatingto a value indicated by the corresponding control data after updating.

It should be noted that any combination of the components describedabove, any component of the present invention, or any manifestationthereof, may be mutually substituted between a method, apparatus,system, and so forth, which are also effective as an embodiment of thepresent invention.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments will now be described, by way of example only, withreference to the accompanying drawings which are meant to be exemplary,not limiting, and wherein like elements are numbered alike in severalFigures, in which:

FIG. 1 is a block diagram showing a lamp system including an ADBfunction;

FIG. 2 is a block diagram showing a lamp system including an automotivelamp according to an embodiment 1;

FIG. 3 is an operation waveform diagram showing the operation of theautomotive lamp shown in FIG. 2 ;

FIG. 4A is a waveform diagram showing a cathode voltage V_(LED) in thelamp system shown in FIG. 2 , and FIG. 4B is a waveform diagram showingthe cathode voltage V_(LED) according to a comparison technique;

FIG. 5 is a circuit diagram showing a converter controller according toan example 1.1;

FIG. 6 is a circuit diagram showing a converter controller according toan example 1.2;

FIG. 7 is a circuit diagram showing a converter controller according toan example 1.3;

FIG. 8 is a circuit diagram showing a converter controller according toan example 1.4;

FIG. 9 is a circuit diagram showing a converter controller according toan example 1.5;

FIG. 10 is a circuit diagram showing a converter controller according toan example 1.6;

FIG. 11 is a circuit diagram showing a specific configuration of theconverter controller shown in FIG. 10 ;

FIG. 12 is a circuit diagram showing a modification of an on signalgenerating circuit;

FIGS. 13A through 13C are circuit diagrams each showing an exampleconfiguration of a current source;

FIG. 14A through 14C are diagrams for explaining a reduction in theswitching frequency in a light load state;

FIG. 15 is a block diagram showing an automotive lamp according to anembodiment 2;

FIG. 16 is an operation waveform diagram showing the operation of theautomotive lamp shown in FIG. 15 ;

FIG. 17 is a block diagram showing an automotive lamp according to anembodiment 3;

FIG. 18 is a block diagram showing an automotive lamp according to anembodiment 4;

FIG. 19 is an operation waveform diagram showing the operation of theautomotive lamp shown in FIG. 18 ;

FIG. 20 is a circuit diagram showing a lighting circuit according to anembodiment 5;

FIG. 21 is a circuit diagram showing a current driver IC and aperipheral circuit thereof according to an embodiment;

FIG. 22 is an operation waveform diagram showing the operation of thecurrent driver IC;

FIG. 23 shows a plan view and a cross-sectional view of a light sourcewith an integrated driver;

FIG. 24 is a circuit diagram showing an automotive lamp according to amodification 1;

FIG. 25 is a block diagram showing a lamp system including an automotivelamp according to an embodiment 6;

FIG. 26 is an operation waveform diagram showing the operation of theautomotive lamp shown in FIG. 25 ;

FIG. 27 is a schematic diagram showing the IV characteristics of aMOSFET and the transition of the operating point of a series transistor;

FIG. 28 is a circuit diagram showing a converter controller according toan example 6.1;

FIG. 29 is a circuit diagram showing a converter controller according toan example 6.2;

FIG. 30 is a circuit diagram showing a converter controller according toan example 6.3;

FIG. 31 is a circuit diagram showing a converter controller according toan example 6.4;

FIG. 32 is a circuit diagram showing a converter controller according toan example 6.5;

FIG. 33 is a circuit diagram showing a converter controller according toan example 6.6;

FIG. 34 is a circuit diagram showing a specific configuration of theconverter controller shown in FIG. 33 ;

FIG. 35 is a circuit diagram showing a current source according to amodification 6.2;

FIG. 36A through 36C are circuit diagrams each showing a modification ofthe on signal generating circuit;

FIG. 37 is a circuit diagram showing a current driver IC and aperipheral circuit thereof according to an embodiment 7;

FIG. 38 is an operation waveform diagram showing the operation of thecurrent driver IC shown in FIG. 37 ;

FIG. 39 shows a plan view and a cross-sectional view of a light sourcewith an integrated driver;

FIGS. 40A through 40C are diagrams for explaining a reduction in theswitching frequency in a light load state;

FIG. 41 is a block diagram showing an automotive lamp according to anembodiment 8;

FIG. 42 is a block diagram showing an automotive lamp according to anembodiment 9;

FIG. 43 is an operation waveform diagram showing the operation of theautomotive lamp shown in FIG. 42 ;

FIG. 44 is a circuit diagram showing a lighting circuit according to anembodiment 10; and

FIG. 45 is a circuit diagram showing an automotive lamp according to amodification.

DETAILED DESCRIPTION OF THE INVENTION

Description will be made below regarding the present invention based onpreferred embodiments with reference to the drawings. The same orsimilar components, members, and processes are denoted by the samereference numerals, and redundant description thereof will be omitted asappropriate. The embodiments have been described for exemplary purposesonly, and are by no means intended to restrict the present invention.Also, it is not necessarily essential for the present invention that allthe features or a combination thereof be provided as described in theembodiments.

In the present specification, the state represented by the phrase “themember A is coupled to the member B” includes a state in which themember A is indirectly coupled to the member B via another member thatdoes not substantially affect the electric connection between them, orthat does not damage the functions or effects of the connection betweenthem, in addition to a state in which they are physically and directlycoupled.

Similarly, the state represented by the phrase “the member C is providedbetween the member A and the member B” includes a state in which themember A is indirectly coupled to the member C, or the member B isindirectly coupled to the member C via another member that does notsubstantially affect the electric connection between them, or that doesnot damage the functions or effects of the connection between them, inaddition to a state in which they are directly coupled.

In the present specification, the reference symbols denoting electricsignals such as a voltage signal, current signal, or the like, and thereference symbols denoting circuit elements such as a resistor,capacitor, or the like, also represent the corresponding voltage value,current value, resistance value, or capacitance value as necessary.

Overview of the Embodiments 1 Through 5

An embodiment of the present invention disclosed in the presentspecification relates to a lighting circuit structured to be capable ofturning on multiple semiconductor light sources. The lighting circuitincludes: multiple current sources each of which is to be coupled to acorresponding semiconductor light source; a switching converterstructured to supply a driving voltage across each of multiple seriesconnection circuits each formed of the semiconductor light source andthe current source; and a converter controller employing a ripplecontrol method. The converter controller turns on the switchingtransistor of the switching converter in response to a voltage acrossany one of the multiple current sources decreasing to a bottom limitvoltage.

The bottom limit voltage is maintained at the minimum level that ensuresthat each current source is able to generate a predetermined drivingcurrent. This arrangement allows power loss to be reduced for eachcurrent source.

Also, the converter controller may turn off the switching transistorafter the on time elapses after the switching transistor is turned on.

Also, the on time may be feedback controlled such that the switchingfrequency of the switching transistor approaches a target frequency.

Also, the converter controller may turn off the switching transistor inresponse to the driving voltage reaching an upper limit voltage.

Also, the upper limit voltage may be feedback controlled such that theswitching frequency of the switching transistor approaches a targetfrequency.

Also, the multiple current sources may each be structured to allow theon/off state thereof to be controlled independently. Also, the bottomlimit voltage may be raised according to a reduction in the number ofon-state current sources from among the multiple current sources. Thisarrangement is capable of preventing the switching frequency frombecoming excessively low in a light load state. In a case in which thebottom limit voltage is raised, this involves an increase in heatgeneration in each current source. However, there is only a small numberof on-state current sources. Accordingly, the increase in the sum totalof the heat generation does not become a problem.

Also, the multiple current sources may each be structured to allow theon/off state thereof to be controlled independently. Also, the targetfrequency may be changed according to the number of on-state currentsources from among the multiple current sources.

Also, the multiple current sources may each be structured to allow theon/off state thereof to be controlled independently. Also, the lightingcircuit may further include a dummy load coupled to an output of theswitching converter, and structured to be set to an enable stateaccording to the number of on-state current sources from among themultiple current sources. By operating the dummy load in the light loadstate, this arrangement is capable of suppressing a reduction in theswitching frequency.

Also, after a predetermined period of time elapses after the switchingtransistor is turned off, the dummy load may reduce the driving voltage.This arrangement allows the switching frequency to be determinedaccording to the predetermined period of time.

Also, when the driving voltage exceeds a predetermined threshold value,the switching transistor may be forcibly turned off.

Also, the multiple semiconductor light sources and the multiple currentsources may be arranged in the form of a module.

With an embodiment, the lighting circuit may be provided to anautomotive lamp.

Embodiments 1 Through 5 Embodiment 1

FIG. 2 is a block diagram showing a lamp system 1 including anautomotive lamp 100 according to an embodiment 1. The lamp system 1includes a battery 2, an in-vehicle ECU (Electronic Control Unit) 4, andan automotive lamp 100. The automotive lamp 100 is configured as avariable light distribution headlamp having an ADB function. Theautomotive lamp 100 generates a light distribution according to acontrol signal received from the in-vehicle ECU 4.

The automotive lamp 100 includes multiple (N 2) semiconductor lightsources 102_1 through 102_N, a lamp ECU 110, and a lighting circuit 200.Each semiconductor light source 102 may preferably be configured usingan LED. Also, various kinds of light-emitting elements such as an LD,organic EL, or the like, may be employed. Each semiconductor lightsource 102 may include multiple light-emitting elements coupled inseries and/or coupled in parallel. It should be noted that the number ofchannels, i.e., N, is not restricted in particular. Also, N may be 1.

The lamp ECU 110 includes a switch 112 and a microcontroller 114. Themicrocontroller (processor) 114 is coupled to the in-vehicle ECU 4 via abus such as a CAN (Controller Area Network) or LIN (Local InterconnectNetwork) or the like. This allows the microcontroller 114 to receivevarious kinds of information such as a turn-on/turn-off instruction,etc. The microcontroller 114 turns on the switch 112 according to aturn-on instruction received from the in-vehicle ECU 4. In this state, apower supply voltage (battery voltage V_(BAT)) is supplied from thebattery 2 to the lighting circuit 200.

Furthermore, the microcontroller 114 receives a control signal forindicating the light distribution pattern from the in-vehicle ECU 4, andcontrols the lighting circuit 200. Also, the microcontroller 114 mayreceive information that indicates the situation ahead of the vehiclefrom the in-vehicle ECU 4, and may autonomously generate the lightdistribution pattern based on the information thus received.

The lighting circuit 200 supplies the driving currents I_(LED1) throughI_(LEDN) to the multiple semiconductor light sources 102_1 through 102_Nso as to provide a desired light distribution pattern.

The lighting circuit 200 includes multiple current sources 210_1 through210_N, a switching converter 220, and a converter controller 230. Eachcurrent source 210_i (i=1, 2, . . . , N) is coupled to the correspondingsemiconductor light source 102_i in series. The current source 210_ifunctions as a constant current driver that stabilizes the drivingcurrent I_(LEDi) that flows through the semiconductor light source 102_ito a predetermined current amount.

The multiple current sources 210_1 through 210_N are each configured tobe capable of controlling their on/off states independently according toPWM signals S_(PWM1) through S_(PWMN) generated by the lightdistribution controller 116. When the PWM signal S_(PWMi) is set to theon level (e.g., high level), the driving current I_(LEDi) flows, therebyturning on the semiconductor light source 102_i. Conversely, when thePWM signal S_(PWMi) is set to the off level (e.g., low level), thedriving current I_(LEDi) is set to zero, thereby turning off thesemiconductor light source 102_i. By changing the duty cycle of the PWMsignal S_(PWM1), such an arrangement allows the effective luminance ofthe semiconductor light source 102_i to be changed (PWM dimming).

The switching converter 220 supplies a driving voltage V_(OUT) across aseries connection circuit of the semiconductor light source 102 and thecurrent source 210. The switching converter 220 is configured as astep-down converter (Buck converter) including a switching transistorM₁, a rectification diode D₁, an inductor L₁, and an output capacitorC₁.

The converter controller 230 controls the switching converter 220 usinga ripple control method. More specifically, the converter controller 230turns on the switching transistor M₁ of the switching converter 220 whenthe voltage across any one of the multiple current sources 210, i.e.,the voltage V_(LED) at connection nodes that couple any one from amongthe current sources 210 and the corresponding semiconductor light source102, decreases to a predetermined bottom limit voltage V_(BOTTOM).

Furthermore, when a predetermined turn-off condition is satisfied, theconverter controller 230 switches a control pulse S₁ to the off level(high level), thereby turning off the switching transistor M₁. Theturn-off condition may be that the output voltage V_(OUT) of theswitching converter 220 has reached a predetermined upper limit voltageV_(UPPER).

The above is the configuration of the automotive lamp 100. Next,description will be made regarding the operation thereof.

FIG. 3 is an operation waveform diagram showing the operation of theautomotive lamp 100 shown in FIG. 2 . For ease of understanding,description will be made regarding an example in which N=3. Furthermore,description will be made assuming that there is only negligible elementvariation between the multiple current sources 210_1 through 210_N.Furthermore, description will be made assuming that the relationV_(F1)>V_(F2)>V_(F3) holds true due to element variation between thesemiconductor light sources 102. For ease of understanding, descriptionwill be made regarding the operation without involving PWM dimming.

In the off period (low-level period in the drawing) of the switchingtransistor M₁, the output capacitor C₁ of the switching converter 220 isdischarged due to a load current I_(OUT) which is the sum total of thedriving currents I_(LED1) through I_(LED3), which lowers the outputvoltage V_(OUT) with time. In actuality, the output capacitor C₁ ischarged or discharged by the difference between the coil current I_(L)that flows through the inductor L₁ and the load current. Accordingly,the increase/decrease of the output voltage V_(OUT) does not necessarilymatch the on/off state of the switching transistor M₁ on the time axis.

The voltages that each occur across each current source 210, i.e., thevoltages (cathode voltages) V_(LED1) through V_(LED3) at the connectionnodes that each connect the corresponding current source 210 and thecorresponding semiconductor light source 102, are represented by thefollowing Expressions.V _(LED1) =V _(OUT) −V _(F1)V _(LED2) =V _(OUT) −V _(E2)V _(LED3) =V _(OUT) −V _(F3)

Accordingly, the voltages V_(LED1) through V_(LED3) each change whilemaintaining a constant voltage difference with respect to the outputvoltage V_(OUT). In this example, the forward voltage V_(F1) at thefirst channel is the largest value. Accordingly, the cathode voltageV_(LED1) at the first channel is the smallest value.

When the cathode voltage V_(LED1) at the first channel decreases to thebottom limit voltage V_(BOTTOM), the switching transistor M₁ is turnedon.

When the switching transistor M₁ is turned on, this raises the coilcurrent I_(L) that flows through the inductor L₁, which switches theoutput voltage V_(OUT) to an increasing phase. Subsequently, when theoutput voltage V_(OUT) reaches the upper limit voltage V_(UPPER), theswitching transistor M₁ is turned off. The lighting circuit 200 repeatsthis operation.

The above is the operation of the lighting circuit 200. The lightingcircuit 200 is capable of maintaining the voltage across each currentsource 210 at a level in the vicinity of the minimum level that ensuresthat each lighting circuit 200 is able to generate predetermined drivingcurrents I_(LED). This arrangement provides reduced power consumption.

As another approach (comparison technique), an arrangement isconceivable in which the cathode voltages V_(LED1) through V_(LEDN) arefeedback controlled using an error amplifier such that the minimumvoltage thereof approaches a predetermined target value V_(REF).

FIG. 4A is a waveform diagram showing the cathode voltage V_(LED)provided by the embodiment. FIG. 4B is a waveform diagram showing thecathode voltage V_(LED) provided by a comparison technique. The cathodevoltages V_(LED) shown in these drawings are each the lowest voltageV_(MIN) from among the multiple cathode voltages.

With the comparison technique, the average of the minimum voltageV_(MIN) from among the cathode voltages V_(LED1) through V_(LEDN)approaches the target voltage V_(REF) by means of the responsecharacteristics of a phase compensation filter provided to a feedbackloop. That is to say, the bottom level V_(MIN_BOTTOM) of the minimumvoltage V_(MIN) is lower than the target voltage V_(REF). In this case,the difference between the bottom level V_(MIN_BOTTOM) and the targetvoltage V_(REF) changes in an unstable manner depending on thesituation. In order to provide stable circuit operation, as indicated bythe solid line in FIG. 4B, there is a need to set V_(REF) to a highvalue assuming that there is a large difference ΔV between the bottomlevel V_(MIN_BOTTOM) and the target voltage V_(REF). However, in asituation in which there is a small difference ΔV′ between them asindicated by the line of alternately long and short dashes, the cathodevoltage V_(LED) is higher than the bottom limit voltage V_(BOTTOM),leading to the occurrence of unnecessary power consumption in thecurrent source. With the embodiment, as shown in FIG. 4A, thisarrangement allows the bottom level of the cathode voltage V_(LED) toapproach the bottom limit voltage V_(BOTTOM), thereby providing furtherreduced power consumption as compared with the comparison technique.

The present invention encompasses various kinds of apparatuses,circuits, and methods that can be regarded as a block configuration or acircuit configuration shown in FIG. 2 , or otherwise that can be derivedfrom the aforementioned description. That is to say, the presentinvention is not restricted to a specific configuration. More specificdescription will be made below regarding an example configuration forclarification and ease of understanding of the essence of the presentinvention and the circuit operation. That is to say, the followingdescription will by no means be intended to restrict the technical scopeof the present invention.

Example 1.1

FIG. 5 is a circuit diagram showing a converter controller 230Faccording to an example 1.1. An on signal generating circuit 240Fincludes multiple comparators 252_1 through 252_N, and a logic gate 254.Each comparator 252_i compares the corresponding cathode voltageV_(LEDi) with the bottom limit voltage V_(BOTTOM). The comparator 252_igenerates a comparison signal that is asserted (e.g., set to the highlevel) when V_(LEDi)<V_(BOTTOM). The logic gate 254 performs a logicaloperation on the outputs (comparison signals) S_(CMP1) through S_(CMPN)of the multiple comparator 252_1 through 252_N. When at least onecomparison signal is asserted, the logic gate 254 asserts the on signalS_(ON). In this example, the logic gate 254 is configured as an OR gate.

An off signal generating circuit 260F generates an off signal S_(OFF)which determines the timing at which the switching transistor M₁ is tobe turned off. A voltage dividing circuit 261 divides the output voltageV_(OUT) such that it is scaled to an appropriate voltage level. Acomparator 262 compares the output voltage V_(OUT)′ thus divided with athreshold value V_(UPPER)′ obtained by scaling the upper limit voltageV_(UPPER). When the relation V_(OUT)>V_(UPPER) is detected, thecomparator 262 asserts the off signal S_(OFF) (e.g., set to the highlevel).

The logic circuit 234 is configured as an SR flip-flop, for example. Thelogic circuit 234 switches its output Q to the on level (e.g., highlevel) in response to the assertion of the on signal S_(ON).Furthermore, the logic circuit 234 switches its output Q to the offlevel (e.g., low level) in response to the assertion of the off signalS_(OFF). It should be noted that the logic circuit 234 is preferablyconfigured as a reset-priority flip-flop in order to set the switchingconverter to a safer state (i.e., off state of the switching transistorM₁) when the assertion of the on signal S_(ON) and the assertion of theoff signal S_(OFF) occur at the same time.

A driver 232 drives the switching transistor M₁ according to the outputQ of the logic circuit 234. As shown in FIG. 2 , in a case in which theswitching transistor M₁ is configured as a P-channel MOSFET, when theoutput Q is set to the on level, the control pulse S₁, which isconfigured as the output of the driver 232, is set to a low voltage(V_(BAT)−V_(G)). When the output Q is set to the off level, the controlpulse S₁ is set to the high voltage (V_(BAT)).

Example 1.2

FIG. 6 is a circuit diagram showing a comparator controller 230Gaccording to an example 1.2. An on signal generating circuit 240Gincludes a minimum value circuit 256 and a comparator 258. The minimumvalue circuit 256 outputs a voltage V_(MIN) that corresponds to theminimum value from among the multiple cathode voltages V_(LED1) throughV_(LEDN). The minimum value circuit 256 may preferably be configuredusing known techniques. The comparator 258 compares the voltage V_(MIN)with a threshold value V_(BOTTOM)′ that corresponds to the bottom limitvoltage V_(BOTTOM). When the relation V_(MIN)<V_(BOTTOM)′ holds true,the comparator 258 asserts the on signal S_(ON) (e.g., set to the highlevel).

With the example 1.1, in a case in which there are a large number ofchannels, the circuit area required by the comparator group is large andthe chip size becomes large. In contrast, with the example 1.2, such anarrangement requires only a single comparator, thereby allowing thecircuit area to be reduced.

Example 1.3

In-vehicle devices are configured to avoid electromagnetic noise bands,i.e., the LW band of 150 kHz to 280 kHz, the AM band of 510 kHz to 1710kHz, and the SW band of 2.8 MHz to 23 MHz. Accordingly, the switchingfrequency of the switching transistor M₁ is preferably stabilized to avalue on the order of 300 kHz to 450 kHz between the LW band and the AMband.

FIG. 7 is a circuit diagram showing a converter controller 230Haccording to an example 1.3. With this example, the upper limit voltageV_(UPPER) is feedback controlled so as to maintain the switchingfrequency of the switching transistor M₁ at a constant value.

An off signal generating circuit 260H includes a frequency detectioncircuit 264 and an error amplifier 266 in addition to the comparator262. The frequency detection circuit 264 monitors the output Q of thelogic circuit 234 or the control pulse S₁, and generates a frequencydetection signal V_(FREQ) that indicates the switching frequency. Theerror amplifier 266 amplifies the difference between the frequencydetection signal V_(FREQ) and the reference voltage V_(FREQ(REF)) thatdefines a target value of the switching frequency (target frequency),and generates the upper limit voltage V_(UPPER) that corresponds to thedifference thus amplified.

With the example 1.3, this arrangement is capable of stabilizing theswitching frequency to a target value. This allows the noisecountermeasures to be provided in a simple manner.

Example 1.4

FIG. 8 is a circuit diagram showing a converter controller 230Iaccording to an example 1.4. The converter controller 230I may turn offthe switching transistor M₁ after the on time T_(ON) elapses after theswitching transistor M₁ is turned on. That is to say, as the turn-offcondition, a condition that the on time T_(ON) elapses after theswitching transistor M₁ is turned off may be employed.

An off signal generating circuit 260I includes a timer circuit 268. Thetimer circuit 268 starts the measurement of the predetermined on timeT_(ON) in response to the on signal S_(ON). After the on time T_(ON)elapses, the timer circuit 268 asserts (e.g., sets to the high level)the off signal S_(OFF). The timer circuit 268 may be configured as amonostable multivibrator (one-shot pulse generator), for example. Also,the timer circuit 268 may be configured as a digital counter or ananalog timer. In order to detect the timing at which the switchingtransistor M₁ is turned on, the timer circuit 268 may receive the outputQ of the logic circuit 234 or the control pulse S₁ as its input signalinstead of the on signal S_(ON).

Example 1.5

FIG. 9 is a circuit diagram showing a converter controller 230Jaccording to an example 1.5. As with the example 1.4, the convertercontroller 230J turns off the switching transistor M₁ after the on timeT_(ON) elapses after the switching transistor M₁ is turned on. An ORgate 241 corresponds to the on signal generating circuit, and generatesthe on signal S_(ON). The timer circuit 268 is configured as amonostable multivibrator or the like. The timer circuit 268 generatesthe pulse signal S_(P) that is set to the high level for a predeterminedon time T_(ON) after the assertion of the on signal S_(ON), and suppliesthe pulse signal S_(P) to the driver 232. It should be noted that,giving consideration to a situation in which the voltages V_(G1) throughV_(GN) are each lower than the threshold value of the OR gate 241 in thestartup operation or the like, an OR gate 231 is provided as anadditional component. With such an arrangement, the logical OR S_(P)′ ofthe on signal S_(ON) and the output S_(P) of the timer circuit 268 issupplied to the driver 232.

Example 1.6

FIG. 10 is a circuit diagram showing a converter controller 230Kaccording to an example 1.6. An off signal generating circuit 260Kfeedback controls the on time T_(ON) so as to maintain the switchingfrequency at a constant value. A variable timer circuit 270 isconfigured as a monostable multivibrator that generates the pulse signalS_(P) that is set to the high level during a period of the on timeT_(ON) after the assertion of the on signal S_(ON). The variable timercircuit 270 is configured to change the on time T_(ON) according to acontrol voltage V_(CTRL).

For example, the variable timer circuit 270 may include a capacitor, acurrent source that charges the capacitor, and a comparator thatcompares the voltage across the capacitor with a threshold value. Thevariable timer circuit 270 is configured such that at least one fromamong the current amount generated by the current source and thethreshold value can be changed according to the control voltageV_(CTRL).

The frequency detection circuit 272 monitors the output Q of the logiccircuit 234 or the control pulse S₁, and generates a frequency detectionsignal V_(FREQ) that indicates the switching frequency. An erroramplifier 274 amplifiers the difference between the frequency detectionsignal V_(FREQ) and the reference voltage V_(FREQ(REF)) that defines atarget value of the switching frequency (target frequency), andgenerates the control voltage V_(CTRL) that corresponds to thedifference thus amplified.

With the example 1.6, this arrangement is capable of stabilizing theswitching frequency to the target value, thereby allowing the noisecountermeasures to be provided in a simple manner.

FIG. 11 is a circuit diagram showing a specific configuration of theconverter control circuit 230K shown in FIG. 10 . Description will bemade regarding the operation of the frequency detection circuit 272. Acombination of a capacitor C₁₁ and a resistor R₁₁ functions as ahigh-pass filter, which can be regarded as a differentiating circuitthat differentiates the output of the OR gate 231 (or the control pulseS₁). Such a high-pass filter can also be regarded as an edge detectioncircuit that detects an edge of the pulse signal S_(P)′. When the outputof the high-pass filter exceeds a threshold value, i.e., when a positiveedge occurs in the pulse signal S_(P)′, a transistor Tr₁₁ turns on so asto discharge the capacitor C₁₂. During the off period of the transistorTr₁₁, the capacitor C₁₂ is charged via a resistor R₁₂. The voltageV_(C12) across the capacitor C₁₂ is configured as a ramp wave insynchronization with the pulse signal S_(P)′. The time length of theslope portion thereof, and the wave height that corresponds to the timelength of the slope portion, change according to the period of the pulsesignal S_(P)′.

A combination of the transistors Tr₁₂ and Tr₁₃, the resistors R₁₃ andR₁₄, and a capacitor C₁₃ is configured as a peak hold circuit. The peakhold circuit holds the peak value of the voltage V_(C12) across thecapacitor C₁₂. The output V_(FREQ) of the peak hold circuit has acorrelation with the period of the pulse signal S_(P)′, i.e., thefrequency thereof.

A comparator COMP1 compares the frequency detection signal V_(FREQ) withthe reference signal V_(FREQ(REF)) that indicates the target frequency.A combination of a resistor R₁₅ and a capacitor C₁₄ is configured as alow-pass filter. The low-pass filter smooths the output of thecomparator COMP1 so as to generate the control voltage V_(CTRL). Thecontrol signal V_(CTRL) is output via a buffer BUF1.

Description will be made regarding the variable timer circuit 270. Theon signal S_(ON) is inverted by an inverter 273. When the inverted onsignal #S_(ON) becomes lower than a threshold value V_(TH1), i.e., whenthe on signal S_(ON) is set to the high level, the output of acomparator COMP2 is set to the high level. This sets a flip-flop SREF,thereby setting the pulse signal S_(P) to the high level.

During the high-level period of the pulse signal S_(P), the transistorM₂₁ is turned off. During the off period of the transistor M₂₁, acurrent source 271 generates a variable current I_(VAR) that correspondsto the control voltage V_(CTRL) so as to charge a capacitor C₁₅. Whenthe voltage V_(C15) across the capacitor C₁₅ reaches a threshold valueV_(TH2), the output of the comparator COMP3 is set to the high level.This resets the flip-flop SREF, thereby switching the pulse signal S_(P)to the low level. As a result, the transistor M₂₁ is turned on, therebyinitializing the voltage V_(C15) of the capacitor C₁₅.

FIG. 12 is a circuit diagram showing a modification of the on signalgenerating circuit 240. In a case in which the comparator 252 isemployed as shown in FIG. 5 , this arrangement supports high-precisionvoltage comparison. However, such an arrangement has a tradeoff problemof a large circuit area and high costs. In order to solve such aproblem, as shown in FIG. 12 , a voltage comparison unit having a simpleconfiguration including a transistor may be employed. A voltagecomparison unit 253 includes a source follower 255 including a PNPbipolar transistor Tr₂₁ and a comparison circuit 257. The output(V_(LED)+V_(BE)) of the source follower 255 configured as an upstreamstage is voltage divided by means of resistors R₂₁ and R₂₂, and theninput to the base of a transistor Tr₂₂. When the voltage V_(LED) to bemonitored decreases, the base voltage of the transistor Tr₂₂ decreases.When the base voltage becomes lower than the on voltage of the bipolartransistor, the current that flows through the transistor Tr₂₂ is cutoff, which sets the output of the voltage comparison unit 253 to thehigh level.

FIG. 12 shows an example in which the outputs of the multiple voltagecomparison units 253 are input to the OR gate 254. However, the presentinvention is not restricted to such an example. Also, such an OR gate254 may be omitted. With such an arrangement, the collectors of thetransistors Tr₂₂ of the multiple voltage comparison units 253 may becoupled so as to form a common collector. Also, a common resistor may beprovided between the common collector and the power supply line V_(cc).

FIGS. 13A through 13C are circuit diagrams each showing an exampleconfiguration of the current source 210. The current source 210 shown inFIG. 13A includes a series transistor M₂, a sensing resistor R_(S), andan error amplifier 212. The series transistor M₂ and the sensingresistor R_(S) are provided in series on a path of the driving currentI_(LEDi). The error amplifier 212 adjusts the voltage V_(G) at a controlelectrode (gate in this example) of the series transistor M₂ such thatthe voltage drop V_(CS) that occurs across the sensing resistor R_(S)approaches a target voltage V_(ADIM). In this example, the seriestransistor M₂ is configured as an N-type (N-channel) MOS transistor. Theerror amplifier 212 is arranged such that the reference voltage V_(ADIM)is input to one input thereof (non-inverting input terminal) and suchthat the voltage V_(CS) (voltage drop that occurs across the sensingresistor R_(S)) at a connection node that couples the series transistorM₂ and the sensing resistor R_(S) is input to the other input thereof(inverting input terminal). The error amplifier 212 provides feedbackcontrol such that V_(CS) approaches V_(ADIM), thereby stabilizing thedriving current I_(LED) with I_(LED)=V_(ADIM)/R_(S) as its target value.

The current source 210 further includes a switch (dimming switch) 214for PWM dimming. The dimming switch 214 is controlled according to a PWMsignal S_(PWM) generated by the dimming controller 116. When the dimmingswitch 214 is turned off, the driving current I_(LED) flows through thecurrent source 210. When the dimming switch 214 is turned on, the seriestransistor M₂ is turned off, which disconnects the driving currentI_(LED). The dimming switch 214 is switched at a high speed at a PWMfrequency of 60 Hz or more (preferably, on the order of 200 to 300 Hz).Furthermore, by adjusting the duty cycle of the PWM frequency, thesemiconductor light source 102 is subjected to PWM dimming control.

In the current source 210 shown in FIG. 13B, the series transistor isconfigured as a P-channel MOSFET. The error amplifier 212 is configuredto have a polarity that is the reverse of that shown in FIG. 13A.

In a case of employing the current source 210 shown in FIG. 13A or 13B,the bottom limit voltage V_(BOTTOM) may preferably be determined asrepresented by the following Expression. Here, ΔV represents anappropriate margin.V _(BOTTOM) =R _(S) ×I _(LED) +V _(SAT) +ΔV

The current source 210 shown in FIG. 13C includes a current mirrorcircuit 216 and a reference current source 218. The current mirrorcircuit 216 multiplies the reference current I_(REF) generated by thereference current source 218 by a predetermined coefficient determinedby a mirror ratio, so as to generate the driving current I_(LED). In acase of employing the current source 210 shown in FIG. 13C, the bottomlimit voltage V_(BOTTOM) may preferably be determined as represented bythe following Expression.V _(BOTTOM) =V _(SAT) +ΔV

Here, V_(SAT) represents the saturation voltage of the current mirrorcircuit, and ΔV represents an appropriate margin.

Embodiment 2

Description has been made in the embodiment 1 regarding an arrangementin which the bottom limit voltage is fixed. In this case, in some cases,such an arrangement has a problem of a reduction in the switchingfrequency in a light load state in which the number of the turned-onlight sources 102 becomes small.

FIGS. 14A through 14C are diagrams for explaining the reduction in theswitching frequency in the light load state. As shown in FIGS. 14A and14B, with the examples shown in FIGS. 7 and 8 , the on time T_(ON) orthe upper limit V_(UPPER) of the output voltage V_(OUT) is feedbackcontrolled so as to stabilize the frequency.

However, in a case in which the pulse width of the control pulse S₁ isexcessively narrowed, such an arrangement is not able to turn on theswitching transistor M₁. Accordingly, such an arrangement is not capableof shortening the pulse width of the control pulse S₁ such that it issmaller than a particular minimum pulse width. In other words, in thelight load state, the pulse width of the control pulse S₁ is fixed tothe minimum pulse width (FIG. 14C). The angle of the downward slope ofthe output voltage V_(OUT) corresponds to the load current, i.e., thenumber of the turned-on semiconductor light sources 102. In a state inwhich the number of turned-on semiconductor light sources 102 becomessmall, the slope of the downward slope becomes smaller, which lowers theswitching frequency. Accordingly, even in a case of supporting thefrequency stabilizing control operation, such an arrangement has thepotential to cause a situation in which the switching frequency is setto a value in the LW band.

In order to solve such a problem, with the embodiment 2, the bottomlimit voltage V_(BOTTOM) is dynamically controlled according to the loadstate so as to suppress the reduction in the switching frequency.

FIG. 15 is a block diagram showing an automotive lamp 100M according tothe embodiment 2. The automotive lamp 100M further includes a bottomlimit voltage setting circuit 280 in addition to the configuration ofthe automotive lamp 100 shown in FIG. 2 . The bottom limit voltagesetting circuit 280 raises the bottom limit voltage V_(BOTTOM) accordingto a reduction in the number of the on-state current sources from amongthe multiple current sources 210. The bottom limit voltage V_(BOTTOM)may be changed in two steps in a stepwise manner. Also, the bottom limitvoltage V_(BOTTOM) may be changed in three or more steps in a stepwisemanner.

For example, the bottom limit voltage setting circuit 280 may judge thenumber of the turned-on light sources based on the PWM signals S_(PWM1)through S_(PWMN) generated by the light distribution controller 116.Also, the bottom limit voltage setting circuit 280 receives, from themicrocontroller 114, a signal that indicates the number of the turned-onlight sources or an instruction value that indicates the bottom limitvoltage V_(BOTTOM) determined based on the number of the turned-on lightsources. Also, with an arrangement described later with reference toFIG. 21 , the number of the turned-on light sources may be judged basedon a signal received by an interface circuit 320.

The configuration of the converter controller 230 is not restricted inparticular. That is to say, the converter controller 230 may have anyone from among the configurations described above.

FIG. 16 is an operation waveform diagram showing the operation of theautomotive lamp 100M shown in FIG. 15 . When the number of the turned-onlight sources becomes small, and accordingly, when the load currentbecomes small, the downward slope of the output voltage V_(OUT) becomesflat. The bottom limit voltage V_(BOTTOM) is raised so as to raise thelower limit voltage of the output voltage V_(OUT) according to thereduction of the slope angle, thereby suppressing an increase in the offtime T_(OFF).

In the light load state, this arrangement is capable of preventing theswitching frequency from becoming excessively low. It should be notedthat, in a case in which the bottom limit voltage V_(BOTTOM) is raised,this involves an increase in heat generation in the current source 210.However, the number of the on-state current sources 210 becomes smaller.Accordingly, the increase in the sum total of the heat generation doesnot become a problem. Description has been made with reference to FIG.16 regarding an arrangement in which the bottom limit voltage V_(BOTTOM)is changed such that the switching frequency is maintained at asubstantially constant level. However, the present invention is notrestricted to such an arrangement. Also, such an arrangement may involvethe change in the switching frequency so long as the switching frequencyis set to a value outside the band that causes a noise problem.

Embodiment 3

FIG. 17 is a block diagram showing an automotive lamp 100N according toan embodiment 3. The automotive lamp 100N further includes a frequencysetting circuit 290 in addition to the configuration of the automotivelamp 100 shown in FIG. 2 . In this embodiment, the converter controller230 is provided with a frequency stabilizing function. Accordingly, theconverter controller 230 may be configured as the converter controller230H shown in FIG. 7 or the converter controller 230J shown in FIG. 10 .

The frequency setting circuit 290 changes the target frequency accordingto the number of the on-state current sources (the number of turned-onlight sources) from among the multiple current sources 210. Morespecifically, when the number of the on-state current sources becomessmaller than a predetermined threshold value, judgment is made that thelight load state has been detected. In this state, the frequency settingcircuit 290 sets the target frequency to a different frequency valuethat is lower than the original target frequency and does not belong toa particular band defined as an electromagnetic noise band. In a case inwhich, in the normal state, the target frequency is set to a frequencyvalue of 300 kHz to 450 kHz between the LW band and AM band, when theoperating state becomes the light load state, the target frequency maypreferably be set to a band (e.g., 100 kHz) that is lower than the LWband and that is higher than the audible band.

With an arrangement shown in FIG. 7 or 10 , the target frequency isdetermined based on the reference voltage V_(FREQ(REF)). Accordingly, ina state in which the number of the turned-on light sources is smallerthan a predetermined threshold, the frequency setting circuit 290 maypreferably reduce the reference voltage V_(FREQ(REF)).

With the embodiment 3, when the frequency is lowered in the light loadstate, such an arrangement is capable of maintaining the frequency suchthat it is outside the frequency range that causes an electromagneticnoise problem that is to be avoided.

Embodiment 4

FIG. 18 is a block diagram showing an automotive lamp 100O according anembodiment 4. The automotive lamp 100O further includes a dummy load 292and a dummy load control circuit 294 in addition to the configuration ofthe automotive lamp 100 shown in FIG. 2 .

The dummy load 292 is coupled to the output of the switching converter220. In the enable state, the dummy load 292 discharges the capacitor C₁of the switching converter 220 so as to lower the output voltageV_(OUT). The dummy load control circuit 294 controls the enable/disablestate of the dummy load 292 based on the number of the on-state currentsources from among the multiple current sources.

The dummy load 292 includes a switch configured as a transistor arrangedbetween the output of the switching converter 220 and the ground. Aftera predetermined time τ elapses from the turning-off of the switchingtransistor M₁, the dummy load control circuit 294 asserts (sets to thehigh level, for example) the enable signal EN, so as to turn on theswitch of the dummy load 292.

FIG. 19 is an operation waveform diagram showing the operation of theautomotive lamp 100O shown in FIG. 18 . When the operating state becomesthe light load state, the enable signal EN is asserted for each cycle,which immediately decreases the output voltage V_(OUT). Subsequently,when the output voltage V_(OUT) decreases to a voltage level thatcorresponds to the bottom limit voltage V_(BOTTOM), the control pulse S₁is set to the high level. That is to say, the upper limit of the offtime T_(OFF) of the switching transistor M₁ is limited by thepredetermined period τ. This restricts the reduction in the switchingfrequency in the light load state.

The dummy load 292 may be configured as a constant current source thatis capable of switching its state between the on state and the offstate. Also, the dummy load 292 may be configured as a combination ofswitches and resistors.

Embodiment 5

Description will be made with reference to FIG. 2 . Typically, there isa tradeoff relation between the on resistance and the breakdown voltageof a transistor. When overshoot occurs in the output voltage V_(OUT) ofthe switching converter, this raise the voltage applied to a transistorthat forms each current source 210. Accordingly, there is a need toconfigure each current source 210 using a high-breakdown-voltageelement. However, such a high-breakdown-voltage element has a large onresistance R_(ON). Accordingly, such an arrangement requires the bottomlimit voltage V_(BOTTOM) to be set to a high value. This leads toproblems of large power consumption and large heat generation.

FIG. 20 is a circuit diagram showing a lighting circuit 200P accordingto an embodiment 5. When the driving voltage V_(OUT) exceeds apredetermined threshold value V_(TH), the lighting circuit 200P forciblyturns off the switching transistor M₁. The lighting circuit 200Pincludes resistors R₃₁ and R₃₂, and a voltage comparator 238. Thevoltage comparator 238 compares the driving voltage V_(OUT)′ divided bythe resistors R₃₁ and R₃₂ with a threshold value V_(TH)′, so as todetect the occurrence of an overvoltage state in the driving voltageV_(OUT).

The converter controller 230P includes a pulse modulator 235, a logicgate 233, and a driver 232. The pulse modulator 235 has the sameconfiguration as those of the converter controllers 230F through 230Krespectively shown in FIGS. 7 through 10 except for the driver 232. Thepulse modulator 235 generates the control pulse S₁′. When the output S₂of the voltage comparator 238 indicates the relation V_(OUT)′<V_(TH)′,the logic gate 233 allows the control pulse S₁′ to pass through as itis. Conversely, when the output S₂ of the voltage comparator 238indicates the relation V_(OUT)′>V_(TH)′, the logic gate 233 forciblysets the level of the control pulse S₁′ to a level that turns off theswitching transistor M₁. In this example, the switching transistor M₁ isconfigured as an N-channel MOSFET. When S₁ is set to the low level, theswitching transistor M₁ is set to the off state. When V_(OUT)′>V_(TH)′,the output S₂ of the voltage comparator 238 is set to the low level. Thelogic gate 233 is configured as an AND gate.

With the present embodiment, the current source 210 is configured usinga transistor having a low on resistance, thereby allowing powerconsumption to be reduced. As a tradeoff, such an arrangement involvessuch a transistor having a low breakdown voltage. However, when anovershoot occurs in the output voltage V_(OUT) of the switchingconverter, the switching transistor M₁ is immediately suspended. Such anarrangement is capable of preventing an overvoltage from being appliedto the transistor of the current source (e.g., the transistor M₂ shownin FIGS. 13A and 13B, the output-side transistor of the current mirrorcircuit 216 shown in FIG. 13C).

Integrated-Driver Light Source

Next, description will be made regarding a light source with anintegrated driver. The multiple current sources 210 may be integrated ona single semiconductor chip, which will be referred as a “current driverIC (Integrated Circuit)” hereafter. FIG. 21 is a circuit diagram showinga current driver IC 300 and a peripheral circuit thereof according tothe embodiment. In addition to multiple current sources 310_1 through310_N, the current driver IC 300 includes an interface circuit 320 and adimming pulse generator 330.

The multiple current sources 310_1 through 310_N are configured toswitch independently between the on state and the off state according toPWM signals S_(PWM1) through S_(PWMN), respectively. The current sources310_1 through 310_N are respectively coupled to the correspondingsemiconductor light sources 102_1 through 102_N in series via cathodepins LED1 through LEDN.

The interface circuit 320 receives multiple control data D₁ throughD_(N) from an external microcontroller (processor 114). The kind of theinterface is not restricted in particular. For example, an SPI (SerialPeripheral Interface) or I²C interface may be employed. The multiplecontrol data D₁ through D_(N) respectively indicate the on/off dutycycles of the multiple current sources 310_1 through 310_N, which areupdated at a first time interval T₁. The first time interval T₁ is setto on the order of 20 ms to 200 ms. For example, the first time intervalT₁ is set to 100 ms.

The dimming pulse generator 330 generates the multiple PWM signalsS_(PWM1) through S_(PWMN) for the multiple current sources 310_1 through310_N based on the multiple control data D₁ through D_(N). In theembodiment described with reference to FIG. 2 , the microcontroller 114generates the multiple PWM signals S_(PWM1) through S_(PWN). In theembodiment 2, the current driver IC 300 has a built-in function ofgenerating the multiple PWM signals S_(PWM1) through S_(PWMN).

The duty cycle of the i-th PWM signal S_(PWMi) is gradually changed at asecond time interval T₂ that is shorter than the first time interval T₁from the corresponding control data D_(i) value before updating to theupdated value thereof (which will be referred to as the “gradual-changemode”). The second time interval T₂ is set to a value on the order of 1ms to 10 ms. For example, the second time interval T₂ is set to 5 ms.

The dimming pulse generator 330 is capable of supporting anon-gradual-change mode in addition to the gradual-change mode. In thenon-gradual-change mode, the duty cycle of the i-th PWM signal S_(PWMi)is allowed to be immediately changed from the corresponding control dataD_(i) value before updating to the updated value thereof.

The dimming pulse generator 330 may preferably be configured todynamically switch its mode between the non-gradual-change mode and thegradual-change mode according to the settings received from themicrocontroller 114. Preferably, the dimming pulse generator 330 isconfigured to dynamically switch its mode between the non-gradual-changemode and the gradual-change mode for each channel (for each dimmingpulse). The setting data that indicates the mode may be appended to thecontrol data D_(i).

A part of or the whole of the on signal generating circuit 240 may beintegrated on the current driver IC 300. The part of the on signalgenerating circuit 240 to be integrated may preferably be determinedaccording to the circuit configuration of the on signal generatingcircuit 240, and specifically, may preferably be determined so as toreduce the number of lines that couple the converter controller 230 andthe current driver IC 300. As shown in FIG. 21 , in a case in which theentire on signal generating circuit 240 is integrated on the currentdriver IC 300, such an arrangement requires only a single line betweenthe converter controller 230 and the current driver IC 300, which isused to transmit the on signal S_(ON). On the other hand, in a case ofemploying the on signal generating circuit 240G shown in FIG. 6 , and ina case in which the minimum value circuit 256 is integrated on thecurrent driver IC 300, such an arrangement requires only a single linebetween the converter controller 230 and the current driver IC 300,through which the minimum voltage V_(MIN) propagates.

Next, description will be made regarding the operation of the currentdriver IC 300. FIG. 22 is an operation waveform diagram showing theoperation of the current driver IC 300. Here, description will be madeassuming that the duty cycle of the PWM signal is changed linearly. Forexample, in a case in which T₁=100 ms, and T₂=5 ms, the duty cycle maypreferably be changed in a stepwise manner with 20 steps. With thedifference between the control data value before updating and thecontrol data value after updating as X %, the duty cycle of the PWMsignal is changed in a stepwise manner with steps of ΔY=(ΔX/20)%.

The above is the operation of the current driver IC 300. The advantagesof the current driver IC 300 can be clearly understood in comparisonwith a comparison technique. If the current driver IC 300 does not havethe function of gradually changing the duty cycle, the microcontroller114 must update the control data D₁ through D_(N) that each indicate theduty cycle at the second time interval T₂. In a case in which the numberof channels N of the semiconductor light sources 102 exceeds severaldozen to 100, such an arrangement requires a high-performancemicrocontroller, i.e., a high-cost microcontroller, configured as themicrocontroller 114. Furthermore, such an arrangement requireshigh-speed communication between the microcontroller 114 and the currentdriver IC 300, thereby leading to the occurrence of a noise problem.

In contrast, with the current driver IC 300 according to the embodiment,this arrangement allows the rate at which the microcontroller 114updates the control data D₁ through D_(N) to be reduced. This allows theperformance required for the microcontroller 114 to be reduced.Furthermore, this allows the communication speed between themicrocontroller 114 and the current driver IC 300 to be reduced, therebysolving the noise problem.

The first time interval T₁ may preferably be configured to be variable.In a situation in which there is only a small change in the duty cycle,the first time interval T₁ is increased so as to reduce the datacommunication amount, thereby allowing power consumption and noise to bereduced.

FIG. 22 shows an example in which the duty cycle is changed linearly.Also, the duty cycle may be changed according to a curve function suchas a quadratic function or an exponential function. In a case ofemploying such a quadratic function, this arrangement provides naturaldimming control with less discomfort.

As shown in FIG. 21 , the multiple semiconductor light sources 102_1through 102_N may be integrated on a single semiconductor chip (die)402. Furthermore, the semiconductor chip 402 and the current driver IC300 may be housed in a single package in the form of a module.

FIG. 23 shows a plan view and a cross-sectional view of theintegrated-driver light source 400. The multiple semiconductor lightsources 102 are formed in a matrix on the front face of thesemiconductor chip 402. The back face of the semiconductor chip 402 isprovided with pairs of back-face electrodes A and K that each correspondto a pair of an anode electrode and a cathode electrode of each of themultiple semiconductor light sources 102. In this drawing, only a singleconnection relation is shown for the semiconductor light source 102_1.

The semiconductor chip 402 and the current driver IC 300 aremechanically joined and electrically coupled. The front face of thecurrent driver IC 300 is provided with front-face electrodes 410 (LED1through LEDN in FIG. 21 ) to be respectively coupled to the cathodeelectrodes K of the multiple semiconductor light sources 102 andfront-face electrodes 412 to be respectively coupled to the anodeelectrodes A of the multiple semiconductor light sources 102. Eachfront-face electrode 412 is coupled to a corresponding bump (or pad) 414provided to a package substrate configured as a back face of the currentdriver IC 300. Also, an unshown interposer may be arranged between thesemiconductor chip 402 and the current driver IC 300.

The kind of the package of the integrated-driver light source 400 is notrestricted in particular. As the package of the integrated-driver lightsource 400, a BAG (Ball Grid Array), PGA (Pin Grid Array), LGA (LandGrid Array), QFP (Quad Flat Package), or the like, may be employed.

In a case in which the semiconductor light sources 102 and the currentdriver IC 300 are each configured as a separate module, a countermeasuremay preferably be provided in which a heat dissipation structure or thelike is attached to each module. In contrast, with the integrated-driverlight source 400 as shown in FIG. 23 , there is a need to release thesum total of heat generated by the light sources 102 and the currentsources 210. Accordingly, such an arrangement has the potential torequire a very large heat dissipation structure. However, by employingthe lighting circuit 200 according to the embodiment, this arrangementis capable of suppressing heat generated by the current sources 210.This allows the size of the heat dissipation structure to be attached tothe integrated-driver light source 400 to be reduced.

MODIFICATIONS

Description will be made regarding modifications relating to theembodiments 1 through 5.

Modification 1

Description has been made in the embodiments regarding an arrangement inwhich the current source 210 is configured as a sink circuit, and iscoupled to the cathode of the corresponding semiconductor light source102. However, the present invention is not restricted to such anarrangement. FIG. 24 is a circuit diagram showing an automotive lamp 100according to a modification 1. In this modification, the cathodes of thesemiconductor light sources 102 are coupled so as to form a commoncathode. Furthermore, each current source 210 configured as a sourcecircuit is coupled to the anode side of the corresponding semiconductorlight source 102. Each current source 210 may be configured bygeometrically reversing the configuration shown in any one of FIGS. 13Athrough 13C. With this arrangement, the polarities (P and N) of eachtransistor may preferably be replaced as necessary. The convertercontroller 230 controls the switching converter 220 based on therelation between the voltages V_(CS) each occurs across each currentsource 210 and the bottom limit voltage V_(BOTTOM).

Modification 2

Any transistor such as the series transistor M₂ or the like may beconfigured as a bipolar transistor. In this case, the gate, source, anddrain correspond to the base, emitter, and collector, respectively.

Modification 3

Description has been made in the embodiments regarding an arrangement inwhich the switching transistor M₁ is configured as a P-channel MOSFET.Also, the switching transistor M₁ may be configured as an N-channelMOSFET. In this case, a bootstrap circuit may be provided as anadditional circuit. Instead of such a MOSFET, an IGBT (Insulated GateBipolar Transistor) or a bipolar transistor may be employed.

Overview of the Embodiments 1 Through 5

An embodiment disclosed in the present specification relates to alighting circuit structured to be capable of turning on multiplesemiconductor light sources. The lighting circuit includes: multiplecurrent sources each of which is to be coupled to a correspondingsemiconductor light source, and each of which includes a seriestransistor and a sensing resistor arranged in series with thecorresponding semiconductor light source, and an error amplifierstructured to adjust the voltage at a control electrode of the seriestransistor based on a voltage drop that occurs across the sensingresistor; a switching converter structured to supply a driving voltageacross each of multiple series connection circuits each formed of asemiconductor light source and a current source; and a convertercontroller structured to operate using a ripple control method. Theconverter controller turns on a switching transistor of the switchingconverter in response to the output voltage of the error amplifierincluded in any one of the multiple current sources satisfying apredetermined turn-on condition.

When the driving current generated by the current source deviates fromits target value, a sudden change occurs in the output voltage of theerror amplifier. The switching converter employs a hysteresis controlmethod. Upon detecting such a sudden change, the switching converterimmediately turns on the switching transistor. This allows the voltageacross each current source to be maintained in the vicinity of thesaturation voltage state, and allows the power consumption to bereduced.

Also, the series transistor may be configured as an N-type transistor.When an output voltage of the error amplifier included in any one of themultiple current sources reaches a predetermined threshold value, theconverter controller may turn on the switching transistor.

Also, the series transistor may be configured as an N-type transistor.Also, the converter controller may turn on the switching transistor inresponse to a maximum value from among output voltages of the pluralityof error amplifiers included in the plurality of semiconductor lightsources satisfying a predetermined turn-on condition.

Also, the series transistor may be configured as a P-type transistor.Also, when an output voltage of the error amplifier included in any oneof the multiple current sources becomes lower than a predeterminedthreshold value, the converter controller may turn on the switchingtransistor.

Also, the converter controller may turn off the switching transistor inresponse to the driving voltage reaching an upper limit voltage. Also,the upper limit voltage may be feedback controlled such that theswitching frequency of the switching transistor approaches a targetfrequency.

Also, the converter controller may turn off the switching transistorafter the on time elapses after the switching transistor is turned on.Also, the on time may be feedback controlled such that the switchingfrequency of the switching transistor approaches a target frequency.

The multiple semiconductor light sources and the multiple currentsources may be arranged in the form of a module. In a case in which thesemiconductor light sources and the current sources are arranged in theform of a module, this further increases a need to reduce the heatgeneration. In a case of employing the hysteresis control method basedon the output voltage of the error amplifier, such an arrangementoperates particularly effectively for such a module.

With an embodiment, the lighting circuit may be provided to anautomotive lamp.

Another embodiment of the present invention disclosed in the presentspecification relates to a current driver circuit structured to drivemultiple semiconductor light sources. The current driver circuitincludes: multiple current sources each structured to allow the on/offstate thereof to be controlled independently according to a PWM signal,and each coupled to a corresponding semiconductor light source inseries; an interface circuit structured to receive, at a first timeinterval, multiple control data that indicate an on/off duty cycle forthe multiple current sources; and a dimming pulse generator structuredto generate multiple PWM signals for the multiple current sources, andto gradually change, at a second time interval that is smaller than thefirst time interval, a duty cycle of each of the multiple PWM signalsfrom a value indicated by the corresponding control data before updatingto a value indicated by the corresponding control data after updating.

In a case in which the current driver circuit is provided with anautomatic duty cycle gradual-change function, i.e., an automaticluminance gradual-change function, the processor is not required toupdate the setting value for the duty cycle with a high frequency. Thisallows the data communication amount to be reduced.

With an embodiment, the duty cycle of each of the multiple PWM signalsmay be immediately changed according to settings from a value indicatedby the corresponding control data before updating to a value indicatedby the corresponding control data after updating. For example, in a casein which the current driver circuit is employed in a variable lightdistribution lamp, in some situations, in order to prevent theoccurrence of glare, there is a need to turn off or reduce a particularillumination provided by a particular semiconductor light source. Thisfunction has an advantage in such a situation.

Also, each of the multiple current sources may include: a seriestransistor and a sensing resistor arranged in series with acorresponding semiconductor light source; an error amplifier structuredto adjust a voltage of a control electrode of the series transistorbased on a voltage drop that occurs across the sensing resistor; and aPWM switch arranged between a gate and a source of the seriestransistor.

Embodiments 6 Through 10 Embodiment 6

FIG. 25 is a block diagram showing a lamp system 1 including anautomotive lamp 100 according to an embodiment 6. The lamp system 1includes a battery 2, an in-vehicle ECU (Electronic Control Unit) 4, andan automotive lamp 100. The automotive lamp 100 is configured as avariable light distribution headlamp having an ADB function. Theautomotive lamp 100 generates a light distribution according to acontrol signal received from the in-vehicle ECU 4.

The automotive lamp 100 includes multiple (N≥2) semiconductor lightsources 102_1 through 102_N, a lamp ECU 110, and a lighting circuit 200.Each semiconductor light source 102 may preferably be configured usingan LED. Also, various kinds of light-emitting elements such as an LD,organic EL, or the like, may be employed. Each semiconductor lightsource 102 may include multiple light-emitting elements coupled inseries and/or coupled in parallel. It should be noted that the number ofchannels, i.e., N, is not restricted in particular. Also, N may be 1.

The lamp ECU 110 includes a switch 112 and a microcontroller 114. Themicrocontroller (processor) 114 is coupled to the in-vehicle ECU 4 via abus such as a CAN (Controller Area Network) or LIN (Local InterconnectNetwork) or the like. This allows the microcontroller 114 to receivevarious kinds of information such as a turn-on/turn-off instruction,etc. The microcontroller 114 turns on the switch 112 according to aturn-on instruction received from the in-vehicle ECU 4. In this state, apower supply voltage (battery voltage V_(BAT)) is supplied from thebattery 2 to the lighting circuit 200.

Furthermore, the microcontroller 114 receives a control signal forindicating the light distribution pattern from the in-vehicle ECU 4, andcontrols the lighting circuit 200. Also, the microcontroller 114 mayreceive information that indicates the situation ahead of the vehiclefrom the in-vehicle ECU 4, and may autonomously generate the lightdistribution pattern based on the information thus received.

The lighting circuit 200 supplies the driving currents I_(LED1) throughI_(LEDN) to the multiple semiconductor light sources 102_1 through 102_Nso as to provide a desired light distribution pattern.

The lighting circuit 200 includes multiple current sources 210_1 through210_N, a switching converter 220, and a converter controller 230. Eachcurrent source 210_i (i=1, 2, . . . , N) is coupled to the correspondingsemiconductor light source 102_i in series. The current source 210_ifunctions as a constant current driver that stabilizes the drivingcurrent I_(LEDi) that flows through the semiconductor light source 102_ito a predetermined current amount.

The multiple current sources 210_1 through 210_N have the sameconfiguration. Accordingly, as a representative example, only theconfiguration of the current source 210_1 is shown. Each current source210 includes a series transistor M₂, a sensing resistor R_(S), and anerror amplifier 212. The series transistor M₂ and the sensing resistorR_(S) are arranged in series on a path of the driving current I_(LEDi).The error amplifier 212 adjusts the voltage V_(G) at a control electrode(gate in this example) of the series transistor M₂ such that the voltagedrop V_(CS) that occurs across the sensing resistor R_(S) approaches thetarget voltage V_(ADIM). In this example, the series transistor M₂ isconfigured as an N-type (N-channel) MOSFET. The error amplifier 212 isarranged such that the reference voltage V_(ADIM) is input to one input(non-inverting input terminal) thereof and such that the voltage V_(CS)(voltage drop that occurs across the sensing resistor R_(S)) at aconnection node that couples the series transistor M₂ and the sensingresistor R_(S) is input to the other input (inverting input terminal)thereof. The error amplifier 212 feedback controls V_(CS) such that itapproaches V_(ADIM). This stabilizes the driving current I_(LED) withI_(LED(REF))=V_(ADIM)/R_(S) as its target value.

Each current source 210 further includes a switch (dimming switch) 214for PWM dimming. The dimming switch 214 is controlled according to thePWM signal S_(PWM) generated by the light distribution controller 116.When the dimming switch 214 is turned off, the driving current I_(LED)flows through the current source 210. Conversely, when the dimmingswitch 214 is turned on, the series transistor M₂ is turned off, whichdisconnects the driving current I_(LED). The dimming switch 214 isswitched at a high speed at a PWM frequency of 60 Hz or more(preferably, on the order of 200 to 300 Hz). By adjusting the duty cyclethereof, the semiconductor light source 102 is subjected to PWM dimmingcontrol.

The switching converter 220 supplies a driving voltage V_(OUT) across aseries connection circuit of the semiconductor light source 102 and thecurrent source 210. The switching converter 220 is configured as astep-down converter (Buck converter) including a switching transistorM₁, a rectification diode D₁, an inductor L₁, and an output capacitorC₁.

The converter controller 230 controls the switching converter 220 usinga ripple control method. More specifically, the converter controller 230generates a turn-on timing at which the switching transistor M₁ is to beturned on, based on the output voltage V_(G) of the error amplifier 212(i.e., gate voltage of the series transistor M₂). Specifically, inresponse to the output voltage V_(G) of the error amplifier 212satisfying a predetermined turn-on condition, the converter controller230 switches the control pulse S₁ to the on level (low level), therebyturning on the switching transistor M₁.

More specifically, when the output voltage V_(G1) of the error amplifier212 exceeds a predetermined threshold value V_(TH), the convertercontroller 230 turns on the switching transistor M₁. In the presentembodiment, the automotive lamp 100 is configured as a multi-channeldevice. The gate voltages V_(G1) through V_(GN) are monitored for allthe channels. When any one of the multiple current sources 210 satisfiesthe turn-on condition described above, the converter controller 230turns on the switching transistor M₁. Specifically, when the gatevoltage V_(Gj) at any channel, i.e., the j-th channel, exceeds thethreshold value V_(TH) in the off period of the switching transistor M₁,the converter controller 230 turns on the switching transistor M₁.

Furthermore, when a predetermined turn-off condition is satisfied, theconverter controller 230 switches a control pulse S₁ to the off level(high level), thereby turning off the switching transistor M₁. Theturn-off condition may be that the output voltage V_(OUT) of theswitching converter 220 has reached a predetermined upper limit voltageV_(UPPER).

The above is the configuration of the automotive lamp 100. Next,description will be made regarding the operation thereof. FIG. 26 is anoperation waveform diagram showing the operation of the automotive lamp100 shown in FIG. 25 . FIG. 27 is a schematic diagram showing the IVcharacteristics of a MOSFET and the transition of the operating point ofa series transistor M₂. For ease of understanding, description will bemade regarding an example in which N=3. Furthermore, description will bemade assuming that there is only negligible element variation betweenthe multiple current sources 210_1 through 210_N. Furthermore,description will be made assuming that the relation V_(F1)>V_(F2)>V_(F3)holds true due to element variation between the semiconductor lightsources 102. For ease of understanding, description will be maderegarding the operation without involving PWM dimming.

Referring to FIG. 26 , in the off period (low-level period in thedrawing) of the switching transistor M₁, the output capacitor C₁ of theswitching converter 220 is discharged due to a load current I_(OUT)which is the sum total of the driving currents I_(LED1) throughI_(LED3), which lowers the output voltage V_(OUT) with time. Inactuality, the output capacitor C₁ is charged or discharged by thedifference between the coil current I_(L) that flows through theinductor L₁ and the load current I_(OUT). Accordingly, theincrease/decrease of the output voltage V_(OUT) does not necessarilymatch the on/off state of the switching transistor M₁ on the time axis.

The voltages that each occur across each current source 210, i.e., thevoltages (cathode voltages) V_(LED1) through V_(LED3) at the connectionnodes that each connect the corresponding current source 210 and thecorresponding semiconductor light source 102, are represented by thefollowing Expressions.V _(LED1) =V _(OUT) −V _(F2)V _(LED2) =V _(OUT) −V _(F2)V _(LED3) =V _(OUT) −V _(F3)

Accordingly, the voltages V_(LED1) through V_(LED3) each change whilemaintaining a constant voltage difference with respect to the outputvoltage V_(OUT). In this example, the forward voltage V_(F1) at thefirst channel is the largest value. Accordingly, the cathode voltageV_(LED1) at the first channel is the smallest value.

The drain-source voltage V_(DS) of the series transistor M₂ at eachchannel is equal to a voltage obtained by subtracting the voltage dropV_(CS) that occurs across the sensing resistor R_(S) from the cathodevoltage V_(LED).V _(DS1) =V _(LED1) −V _(CS1)V _(DS2) =V _(LED2) −V _(CS2)V _(DS3) =V _(LED3) −V _(CS3)

In a case in which the target values I_(LED(REF)) of the drivingcurrents I_(LED) are equal for all the channels, and in a case in whichthe sensing resistors R_(S) have the same resistance value for all thechannels, the voltage drops V_(CS1) through V_(CS3) are the same for allthe channels. In this case, the first channel exhibits the smallestdrain-source voltage V_(DS1).

The series transistor M₂ may be designed to have an element size so asto operate mainly in its saturation range. In the saturation range, theseries transistor M₂ allows the target current I_(LED(REF)) to flow at apredetermined gate voltage level V₀ without depending on thedrain-source voltage V_(DS). That is to say, in the saturation range,the error amplifier 212 feedback controls the gate voltage V_(G1) suchthat it is set to V₀. As the output voltage V_(OUT) becomes lower, theoperation point moves along the line indicated by the arrow (i) in FIG.27 .

In a case in which the gate-source voltage V_(GS) is maintained at aconstant value, when the drain-source voltage V_(DS1) at the firstchannel becomes lower than a pinch-off voltage V_(P)(=V_(GS)−V_(GS(th))), this leads to reduction in the drain current I_(D)(i.e., driving current I_(LED)) (as indicated by the arrow (ii) in FIG.27 ). The reduction in the driving current I_(D) manifests as areduction in the detection voltage V_(CS1). FIG. 26 shows an expandedview of a very small decrease in the detection voltage V_(CS1). In thisstate, the error amplifier 212 feedback controls the gate voltage V_(G1)so as to adjust it to a higher voltage level V1 (as indicated by thearrow (iii) in FIG. 27 ) such that the detection voltage V_(CS1) thathas decreased approaches the target voltage V_(ADIM). The erroramplifier 212 is configured to have a very high gain. Accordingly, sucha very small decrease in the detection voltage V_(CS1) is converted intoa somewhat large rise of the gate voltage V_(GS). When the rise of thegate voltage V_(G1) is detected as a result of a comparison with thethreshold value V_(TH), the switching transistor M₁ is turned on.

When the switching transistor M₁ is turned on, the coil current I_(L)that flows through the inductor L₁ rises, which leads to an increase inthe output voltage V_(OUT). When the output voltage V_(OUT) rises, thisraises the drain-source voltage V_(DS) of the series transistor M₂. In acase in which the gate voltage V_(GS) is maintained at a constant level,when the drain-source voltage V_(DS) rises in the saturation range, thisraises the drain current I_(D) (as indicated by the arrow (iv) in FIG.27 ). The increase in the drain current I_(D) manifests as a rise of thedetection voltage V_(CS1). The error amplifier 212 feedback controls thegate voltage V_(G1) to be adjusted to a lower voltage level V₀ such thatthe detection voltage V_(CS1) that has risen approaches the targetvoltage V_(ADIM) (as indicated by the arrow (v) in FIG. 27 ). When theoutput voltage V_(OUT) further rises in the on period of the switchingtransistor M₁, the operating point moves along the line indicated by thearrow (vi) in FIG. 27 .

Subsequently, when the output voltage V_(OUT) reaches the upper limitvoltage V_(UPPER), the switching transistor M₁ is turned off. Thelighting circuit 200 repeats this operation.

The above is the operation of the lighting circuit 200. With thelighting circuit 200, the series transistor M₂ is allowed to have itsoperating point in the vicinity of the boundary between the linear rangeand the saturation range. This allows the source-drain voltage V_(DS) ofthe series transistor M₂ to be reduced, thereby allowing unnecessarypower consumption in the series transistor M₂ to be reduced.

Description will be made regarding a case in which the PWM dimmingcontrol is performed. When the turned-off period of the PWM dimmingoccurs, and accordingly, when the dimming switch 214 is turned on, thegate voltage V_(G) changes such that it becomes lower. Accordingly, atthis channel in the turned-off state, the gate voltage V_(G) does notcross the threshold voltage V_(TH). Accordingly, in this state, there isno effect on the turn-on operation of the switching transistor M₁. Thatis to say, such an arrangement does not require special processing toeliminate the turned-off channels from the judgment whether or not theturn-on condition has been satisfied.

The present invention encompasses various kinds of apparatuses,circuits, and methods that can be regarded as a block configuration or acircuit configuration shown in FIG. 25 , or otherwise that can bederived from the aforementioned description. That is to say, the presentinvention is not restricted to a specific configuration. More specificdescription will be made below regarding example configurations andmodifications for clarification and ease of understanding of the essenceof the present invention and the circuit operation. That is to say, thefollowing description will by no means be intended to restrict thetechnical scope of the present invention.

Example 6.1

FIG. 28 is a circuit diagram showing a converter controller 230Aaccording to an example 6.1. The converter controller 230A turns on theswitching transistor M₁ in response to the maximum value from among theoutput voltages V_(G1) through V_(GN) of the multiple channels of erroramplifiers 212 satisfying a predetermined turn-on condition (i.e.,exceeding the threshold voltage V_(TH)).

An on-signal generating circuit 240A generates the on signal S_(ON) thatindicates the timing at which the switching transistor M₁ is to beturned on, based on the multiple gate voltages V_(G1) through V_(GN).The on signal generating circuit 240A includes a maximum value circuit242 and a comparator 244. The maximum value circuit 242 generates avoltage that corresponds to the maximum value from among the multiplegate voltages V_(G1) through V_(GN). For example, the maximum valuecircuit 242 may be configured as a diode OR circuit. The output voltageV_(G)′ of the diode OR circuit is Vf lower than the maximum one fromamong the multiple gate voltages V_(G1) through V_(GN). Here, Vfrepresents the forward voltage of the diode.

The comparator 244 compares the output voltage of the maximum valuecircuit 242 with a threshold value V_(TH)′. The threshold value V_(TH)′may preferably be determined to be Vf lower than the threshold voltageV_(TH) described above. When V_(G)′ exceeds V_(TH)′, i.e., when themaximum gate voltage V_(G) exceeds the threshold voltage V_(TH), the onsignal S_(ON), which is the output of the comparator 244, is asserted(set to the high level, for example).

An off signal generating circuit 260A generates an off signal S_(OFF)which determines the timing at which the switching transistor M₁ is tobe turned off. A voltage dividing circuit 261 divides the output voltageV_(OUT) such that it is scaled to an appropriate voltage level. Acomparator 262 compares the output voltage V_(OUT)′ thus divided with athreshold value V_(UPPER)′ obtained by scaling the upper limit voltageV_(UPPER). When the relation V_(OUT)>V_(UPPER) is detected, thecomparator 262 asserts the off signal S_(OFF) (e.g., set to the highlevel).

The logic circuit 234 is configured as an SR flip-flop, for example. Thelogic circuit 234 switches its output Q to the on level (e.g., highlevel) in response to the assertion of the on signal S_(ON).Furthermore, the logic circuit 234 switches its output Q to the offlevel (e.g., low level) in response to the assertion of the off signalS_(OFF). It should be noted that the logic circuit 234 is preferablyconfigured as a reset-priority flip-flop in order to set the switchingconverter to a safer state (i.e., off state of the switching transistorM₁) when the assertion of the on signal S_(ON) and the assertion of theoff signal S_(OFF) occur at the same time.

A driver 232 drives the switching transistor M₁ according to the outputQ of the logic circuit 234. As shown in FIG. 25 , in a case in which theswitching transistor M₁ is configured as a P-channel MOSFET, when theoutput Q is set to the on level, the control pulse S₁, which isconfigured as the output of the driver 232, is set to a low voltage(V_(BAT)−V_(G)). When the output Q is set to the off level, the controlpulse S₁ is set to the high voltage (V_(BAT)).

With the example 6.1, such an arrangement requires only a singlecomparator 244. This allows the circuit area to be reduced as comparedwith the example 6.2.

Example 6.2

FIG. 29 is a circuit diagram showing a converter controller 230Baccording to an example 6.2. An on signal generating circuit 240Bincludes multiple comparators 246_1 through 246_N and a logic gate 248.Each comparator 246_i compares the corresponding gate voltage V_(Gi)with the threshold voltage V_(TH). The logic gate 248 performs a logicaloperation on the outputs of the multiple comparators 246_1 through246_N, so as to generate the on signal S_(ON). In a case of employing apositive logic system, the logic gate 248 may be configured using an ORgate.

Example 6.3

In-vehicle devices are configured to avoid electromagnetic noise bands,i.e., the LW band of 150 kHz to 280 kHz, the AM band of 510 kHz to 1710kHz, and the SW band of 2.8 MHz to 23 MHz. Accordingly, the switchingfrequency of the switching transistor M₁ is preferably stabilized to avalue on the order of 300 kHz to 450 kHz between the LW band and the AMband.

FIG. 30 is a circuit diagram showing a converter controller 230Caccording to an example 6.3. With this example, the upper limit voltageV_(UPPER) is feedback controlled so as to maintain the switchingfrequency of the switching transistor M₁ at a constant value.

An off signal generating circuit 260C includes a frequency detectioncircuit 264 and an error amplifier 266 in addition to the comparator262. The frequency detection circuit 264 monitors the output Q of thelogic circuit 234 or the control pulse S₁, and generates a frequencydetection signal V_(FREQ) that indicates the switching frequency. Theerror amplifier 266 amplifies the difference between the frequencydetection signal V_(FREQ) and the reference voltage V_(FREQ(REF)) thatdefines a target value of the switching frequency, and generates theupper limit voltage V_(UPPER) that corresponds to the difference thusamplified.

With the example 6.3, this arrangement is capable of stabilizing theswitching frequency to a target value. This allows the noisecountermeasures to be provided in a simple manner.

Example 6.4

FIG. 31 is a circuit diagram showing a converter controller 230Daccording to an example 6.4. The converter controller 230D may turn offthe switching transistor M₁ after the on time T_(ON) elapses after theswitching transistor M₁ is turned on. That is to say, as the turn-offcondition, a condition that the on time T_(ON) elapses after theswitching transistor M₁ is turned off may be employed.

An off signal generating circuit 260D includes a timer circuit 268. Thetimer circuit 268 starts the measurement of the predetermined on timeT_(ON) in response to the on signal S_(ON). After the on time T_(ON)elapses, the timer circuit 268 asserts (e.g., sets to the high level)the off signal S_(OFF). The timer circuit 268 may be configured as amonostable multivibrator (one-shot pulse generator), for example. Also,the timer circuit 268 may be configured as a digital counter or ananalog timer. In order to detect the timing at which the switchingtransistor M₁ is turned on, the timer circuit 268 may receive the outputQ of the logic circuit 234 or the control pulse S₁ as its input signalinstead of the on signal S_(ON).

Example 6.5

FIG. 32 is a circuit diagram showing a converter controller 230Faccording to an example 6.5. As with the example 6.4, the convertercontroller 230F turns off the switching transistor M₁ after the on timeT_(ON) elapses after the switching transistor M₁ is turned on. An ORgate 241 corresponds to the on signal generating circuit, and generatesthe on signal S_(ON). The timer circuit 268 is configured as amonostable multivibrator or the like. The timer circuit 268 generatesthe pulse signal S_(P) that is set to the high level for a predeterminedon time T_(ON) after the assertion of the on signal S_(ON), and suppliesthe pulse signal S_(P) to the driver 232. It should be noted that,giving consideration to a situation in which the voltages V_(G1) throughV_(GN) are each lower than the threshold value of the OR gate 241 in thestartup operation or the like, an OR gate 231 is provided as anadditional component. With such an arrangement, the logical OR S_(P)′ ofthe on signal S_(ON) and the output S_(P) of the timer circuit 268 issupplied to the driver 232.

Example 6.6

FIG. 33 is a circuit diagram showing a converter controller 230Eaccording to an example 6.6. An off signal generating circuit 260Efeedback controls the on time T_(ON) so as to maintain the switchingfrequency at a constant value. A variable timer circuit 270 isconfigured as a monostable multivibrator that generates the pulse signalS_(P) that is set to the high level during a period of the on timeT_(ON) after the assertion of the on signal S_(ON). The variable timercircuit 270 is configured to change the on time T_(ON) according to acontrol voltage V_(CTRL).

For example, the variable timer circuit 270 may include a capacitor, acurrent source that charges the capacitor, and a comparator thatcompares the voltage across the capacitor with a threshold value. Thevariable timer circuit 270 is configured such that at least one fromamong the current amount generated by the current source and thethreshold value can be changed according to the control voltageV_(CTRL).

The frequency detection circuit 272 monitors the output Q of the logiccircuit 234 or the control pulse S₁, and generates a frequency detectionsignal V_(FREQ) that indicates the switching frequency. An erroramplifier 274 amplifiers the difference between the frequency detectionsignal V_(FREQ) and the reference voltage V_(FREQ(REF)) that defines atarget value of the switching frequency, and generates the controlvoltage V_(CTRL) that corresponds to the difference thus amplified.

With the example 6.6, this arrangement is capable of stabilizing theswitching frequency to the target value, thereby allowing the noisecountermeasures to be provided in a simple manner.

FIG. 34 is a circuit diagram showing a specific configuration of theconverter control circuit 230E shown in FIG. 33 . Description will bemade regarding the operation of the frequency detection circuit 272. Acombination of a capacitor C₁₁ and a resistor R₁₁ functions as ahigh-pass filter, which can be regarded as a differentiating circuitthat differentiates the pulse signal S_(P)′ which is the output of theOR gate 231 (or the control pulse S₁). Such a high-pass filter can alsobe regarded as an edge detection circuit that detects an edge of thepulse signal S_(P)′. When the output of the high-pass filter exceeds athreshold value, i.e., when a positive edge occurs in the pulse signalS_(P)′, a transistor Tr₁₁ turns on so as to discharge the capacitor C₁₂.During the off period of the transistor Tr₁₁, the capacitor C₁₂ ischarged via a resistor R₁₂. The voltage V_(C12) across the capacitor C₁₂is configured as a ramp wave in synchronization with the pulse signalS_(P)′. The time length of the slope portion thereof, and the waveheight that corresponds to the time length of the slope portion, changeaccording to the period of the pulse signal S_(P)′.

A combination of the transistors Tr₁₂ and Tr₁₃, the resistors R₁₃ andR₁₄, and a capacitor C₁₃ is configured as a peak hold circuit. The peakhold circuit holds the peak value of the voltage V_(C12) across thecapacitor C₁₂. The output V_(FREQ) of the peak hold circuit has acorrelation with the period of the pulse signal S_(P)′, i.e., thefrequency thereof.

A comparator COMP1 compares the frequency detection signal V_(FREQ) withthe reference signal V_(FREQ(REF)) that indicates the target frequency.A combination of a resistor R₁₅ and a capacitor C₁₄ is configured as alow-pass filter. The low-pass filter smooths the output of thecomparator COMP1 so as to generate the control voltage V_(CTRL). Thecontrol signal V_(CTRL) is output via a buffer BUF1.

Description will be made regarding the variable timer circuit 270. Theon signal S_(ON) is inverted by an inverter 273. When the inverted onsignal #S_(ON) becomes lower than a threshold value V_(TH1), i.e., whenthe on signal S_(ON) is set to the high level, the output of acomparator COMP2 is set to the high level. This sets a flip-flop SREF,thereby setting the pulse signal S_(P) to the high level.

During the high-level period of the pulse signal S_(P), the transistorM₂₁ is turned off. During the off period of the transistor M₂₁, acurrent source 271 generates a variable current I_(VAR) that correspondsto the control voltage V_(CTRL) so as to charge a capacitor C₁₅. Whenthe voltage V_(C15) across the capacitor C₁₅ reaches a threshold valueV_(TH2), the output of the comparator COMP3 is set to the high level.This resets the flip-flop SREF, thereby switching the pulse signal S_(P)to the low level. As a result, the transistor M₂₁ is turned on, therebyinitializing the voltage V_(C15) of the capacitor C₁₅.

Next, description will be made regarding modifications relating to theembodiment 6.

Modification 6.1

Also, as the turn-off condition, the converter controller 230 may employthe drain voltage (cathode voltage of the semiconductor light source102) of the series transistor M₂ for each channel. For example, as theturn-off condition, a condition may be employed in which the maximum (orminimum) from among the cathode voltages of the multiple channels of thesemiconductor light sources 102 reaches an upper limit voltage.

Modification 6.2

Description has been made in the embodiment 6 in which an N-typetransistor is employed as the series transistor M₂ of the current source210. Also, a P-type transistor (P-channel MOSFET) may be employed. FIG.35 is a circuit diagram showing a current source 210 according to amodification 6.2. In this case, when the output voltage V_(OUT)decreases, in order to maintain the driving current I_(LED), feedbackcontrol is applied in a direction in which the gate voltage V_(G) isreduced. Accordingly, as the turn-on condition, a condition may beemployed in which the gate voltage V_(G) becomes lower than apredetermined threshold value at any channel. The dimming switch 214 maybe provided between the gate and the source of the series transistor M₂.

Modification 6.3

Any transistor such as the series transistor M₂ or the like may beconfigured as a bipolar transistor. In this case, the gate, source, anddrain correspond to the base, emitter, and collector, respectively.

Modification 6.4

Description has been made in the embodiment 6 regarding an arrangementin which the switching transistor M₁ is configured as a P-channelMOSFET. Also, the switching transistor M₁ may be configured as anN-channel MOSFET. In this case, a bootstrap circuit may be provided asan additional circuit. Instead of such a MOSFET, an IGBT (Insulated GateBipolar Transistor) or a bipolar transistor may be employed.

Modification 6.5

Description has been made in the embodiment 6 regarding an arrangementin which the output voltage of the error amplifier 212 (gate voltageV_(G) of the series transistor M₂) is directly monitored, and judgmentis made regarding whether or not the output voltage of the erroramplifier 212 thus monitored satisfies the turn-on condition. However,the present invention is not restricted to such an arrangement. Forexample, an internal node of the error amplifier 212 that generates avoltage having a correlation with the output voltage may be monitored.That is to say, the output voltage of the error amplifier 212 may beindirectly monitored.

Modification 6.6

Description has been made in the embodiment 6 regarding an arrangementin which the comparator 244 is used to detect a sudden change in theoutput voltage (gate voltage V_(G)) of the error amplifier 212. However,the present invention is not restricted to such an arrangement. FIGS.36A through 36C are circuit diagrams each showing a modification of theon signal generating circuit 240. As shown in FIG. 36A, instead of thecomparator 244 shown in FIG. 28 , a MOSFET or a bipolar transistor maybe employed as the voltage comparing unit. For example, the outputvoltage V_(G)′ of the maximum value circuit 242 may be divided by aresistor voltage dividing circuit 250. Furthermore, the voltage V_(G)″thus divided may be input to the gate (or base) of the transistor 251.With such an arrangement, the on signal S_(ON) may be generatedaccording to the on/off switching operation of the transistor 251.

FIG. 36B shows a modification of the circuit configuration shown in FIG.29 . Specifically, the comparator 244 is omitted for each channel.Instead, resistor voltage dividing circuits 254_1 through 254_N eachhaving an appropriate voltage diving ratio are provided. The gatevoltages V_(G1)′ through V_(GN)′ thus divided are input to the logicgate 256. In this case, when any one of the gate voltages V_(G)′ thusdivided for each channel exceeds a high/low threshold value of the logicgate 256, the on signal S_(ON) is asserted. FIG. 36C is a circuitdiagram showing an example in which, as the logic gate shown in FIG.36B, a NOR gate is employed.

Embodiment 7

An embodiment 7 relates to a current driver. The multiple currentsources 210 may be integrated on a single semiconductor chip, which willbe referred to as a “current driver IC (Integrated Circuit).” FIG. 37 isa circuit diagram showing a current driver IC 300 and a peripheralcircuit thereof according to the embodiment 7. In addition to multiplecurrent sources 310_1 through 310_N, the current driver IC 300 includesan interface circuit 320 and a dimming pulse generator 330.

As shown in the embodiment 6, the multiple current sources 310_1 through310_N are configured to switch independently between the on state andthe off state according to PWM signals S_(PWM1) through S_(PWMN),respectively. The current sources 310_1 through 310_N are respectivelycoupled to the corresponding semiconductor light sources 102_1 through102_N in series via cathode pins LED1 through LEDN.

The interface circuit 320 receives multiple control data D₁ throughD_(N) from an external microcontroller (processor 114). The kind of theinterface is not restricted in particular. For example, an SPI (SerialPeripheral Interface) or I²C interface may be employed. The multiplecontrol data D₁ through D_(N) respectively indicate the on/off dutycycles of the multiple current sources 310_1 through 310_N, which areupdated at a first time interval T₁. The first time interval T₁ is setto on the order of 20 ms to 200 ms. For example, the first time intervalT₁ is set to 100 ms.

The dimming pulse generator 330 generates the multiple PWM signalsS_(PWM1) through S_(PWMN) for the multiple current sources 310_1 through310_N based on the multiple control data D₁ through D_(N). In theembodiment 6 (FIG. 25 ), the microcontroller 114 generates the multiplePWM signals S_(PWM1) through S_(PWN). In the embodiment 7, the currentdriver IC 300 has a built-in function of generating the multiple PWMsignals S_(PWM1) through S_(PWMN).

The duty cycle of the i-th PWM signal S_(PWMi) is gradually changed at asecond time interval T₂ that is shorter than the first time interval T₁from the corresponding control data D_(i) value before updating to theupdated value thereof (which will be referred to as the “gradual-changemode”). The second time interval T₂ is set to a value on the order of 1ms to 10 ms. For example, the second time interval T₂ is set to 5 ms.

The dimming pulse generator 330 is capable of supporting anon-gradual-change mode in addition to the gradual-change mode. In thenon-gradual-change mode, the duty cycle of the i-th PWM signal S_(PWMi)is allowed to be immediately changed from the corresponding control dataD_(i) value before updating to the updated value thereof.

The dimming pulse generator 330 may preferably be configured todynamically switch its mode between the non-gradual-change mode and thegradual-change mode according to the settings received from themicrocontroller 114. Preferably, the dimming pulse generator 330 isconfigured to dynamically switch its mode between the non-gradual-changemode and the gradual-change mode for each channel (for each dimmingpulse). The setting data that indicates the mode may be appended to thecontrol data D_(i).

In a case in which the switching transistor M₁ is controlled in themanner described in the embodiment 6, a part of or the whole of the onsignal generating circuit 240 may be integrated on the current driver IC300. The part of the on signal generating circuit 240 to be integratedmay preferably be determined according to the circuit configuration ofthe on signal generating circuit 240. Specifically, the part of the onsignal generating circuit 240 to be integrated may preferably determinedso as to reduce the number of lines that couple the converter controller230 and the current driver IC 300. As shown in FIG. 37 , in a case inwhich the maximum value circuit 242, which is a part of the on signalgenerating circuit 240, is integrated on the current driver IC 300, suchan arrangement requires only a single line between the convertercontroller 230 and the current driver IC 300, which is used to transmitthe maximum voltage V_(G)′ from among the multiple gate voltages. In acase in which the whole of the on signal generating circuit 240 isintegrated on the current driver IC 300, such an arrangement requiresonly a single line between the converter controller 230 and the currentdriver IC 300, which is used to transmit the on signal S_(ON).

Next, description will be made regarding the operation of the currentdriver IC 300. FIG. 38 is an operation waveform diagram showing theoperation of the current driver IC 300 shown in FIG. 37 . Here,description will be made assuming that the duty cycle of the PWM signalis changed linearly. For example, in a case in which T₁=100 ms, and T₂=5ms, the duty cycle may preferably be changed in a stepwise manner with20 steps. With the difference between the control data value beforeupdating and the control data value after updating as X %, the dutycycle of the PWM signal is changed in a stepwise manner with steps ofΔY=(ΔX/20)%.

The above is the operation of the current driver IC 300. The advantagesof the current driver IC 300 can be clearly understood in comparisonwith a comparison technique. If the current driver IC 300 does not havethe function of gradually changing the duty cycle, the microcontroller114 must update the control data D₁ through D_(N) that each indicate theduty cycle at the second time interval T₂. In a case in which the numberof channels N of the semiconductor light sources 102 exceeds severaldozen to 100, such an arrangement requires a high-performancemicrocontroller, i.e., a high-cost microcontroller, configured as themicrocontroller 114. Furthermore, such an arrangement requireshigh-speed communication between the microcontroller 114 and the currentdriver IC 300, thereby leading to the occurrence of a noise problem.

In contrast, with the current driver IC 300 according to the embodiment,this arrangement allows the rate at which the microcontroller 114updates the control data D₁ through D_(N) to be reduced. This allows theperformance required for the microcontroller 114 to be reduced.Furthermore, this allows the communication speed between themicrocontroller 114 and the current driver IC 300 to be reduced, therebysolving the noise problem.

The first time interval T₁ may preferably be configured to be variable.In a situation in which there is only a small change in the duty cycle,the first time interval T₁ is increased so as to reduce the datacommunication amount, thereby allowing power consumption and noise to bereduced.

FIG. 38 shows an example in which the duty cycle is changed linearly.Also, the duty cycle may be changed according to a curve function suchas a quadratic function or an exponential function. In a case ofemploying such a quadratic function, this arrangement provides naturaldimming control with less discomfort.

As shown in FIG. 37 , the multiple semiconductor light sources 102_1through 102_N may be integrated on a single semiconductor chip (die)402. Furthermore, the semiconductor chip 402 and the current driver IC300 may be housed in a single package in the form of a module.

FIG. 39 shows a plan view and a cross-sectional view of theintegrated-driver light source 400. The multiple semiconductor lightsources 102 are formed in a matrix on the front face of thesemiconductor chip 402. The back face of the semiconductor chip 402 isprovided with pairs of back-face electrodes A and K that each correspondto a pair of an anode electrode and a cathode electrode of each of themultiple semiconductor light sources 102. In this drawing, only a singleconnection relation is shown in an expanded view of the semiconductorlight source 102_1.

The semiconductor chip 402 and the current driver IC 300 aremechanically joined and electrically coupled. The front face of thecurrent driver IC 300 is provided with front-face electrodes 410 (LED1through LEDN in FIG. 37 ) to be respectively coupled to the cathodeelectrodes K of the multiple semiconductor light sources 102 andfront-face electrodes 412 to be respectively coupled to the anodeelectrodes A of the multiple semiconductor light sources 102. Eachfront-face electrode 412 is coupled to a corresponding bump (or pad) 414provided to a package substrate configured as a back face of the currentdriver IC 300. Also, an unshown interposer may be arranged between thesemiconductor chip 402 and the current driver IC 300.

The kind of the package of the integrated-driver light source 400 is notrestricted in particular. As the package of the integrated-driver lightsource 400, a BAG (Ball Grid Array), PGA (Pin Grid Array), LGA (LandGrid Array), QFP (Quad Flat Package), or the like, may be employed.

In a case in which the semiconductor light sources 102 and the currentdriver IC 300 are each configured as a separate module, a countermeasuremay preferably be provided in which a heat dissipation structure or thelike is attached to each module. In contrast, with the integrated-driverlight source 400 as shown in FIG. 39 , there is a need to release thesum total of heat generated by the light sources 102 and the currentsources 210. Accordingly, such an arrangement has the potential torequire a very large heat dissipation structure. However, by employingthe lighting circuit 200 according to the embodiment, this arrangementis capable of suppressing heat generated by the current sources 210.This allows the size of the heat dissipation structure to be attached tothe integrated-driver light source 400 to be reduced.

Embodiment 8

With the automotive lamp 100 according to the embodiment 6, in somecases, such an arrangement has a problem of a reduction in the switchingfrequency in a light load state in which the number of the turned-onlight sources 102 becomes small.

FIGS. 40A through 40C are diagrams for explaining the reduction in theswitching frequency in the light load state. As shown in FIGS. 37A and37B, with the examples shown in FIGS. 30 and 33 , the on time T_(ON) orthe upper limit V_(UPPER) of the output voltage V_(OUT) is feedbackcontrolled so as to stabilize the frequency.

However, in a case in which the pulse width of the control pulse S₁ isexcessively narrowed, such an arrangement is not able to turn on theswitching transistor M₁. Accordingly, such an arrangement is not capableof shortening the pulse width of the control pulse S₁ such that it issmaller than a particular minimum pulse width. In other words, in thelight load state, the pulse width of the control pulse S₁ is fixed tothe minimum pulse width (FIG. 37C). The angle of the downward slope ofthe output voltage V_(OUT) corresponds to the load current, i.e., thenumber of the turned-on semiconductor light sources 102. In a state inwhich the number of turned-on semiconductor light sources 102 becomessmall, the slope of the downward slope becomes smaller, which lowers theswitching frequency. Accordingly, even in a case of supporting thefrequency stabilizing control operation, such an arrangement has thepotential to cause a situation in which the switching frequency is setto a value in the LW band.

FIG. 41 is a block diagram showing an automotive lamp 100X according toan embodiment 8. The automotive lamp 100X further includes a frequencysetting circuit 290 in addition to the configuration of the automotivelamp 100 shown in FIG. 25 . In this embodiment, the converter controller230 is provided with a frequency stabilizing function. Accordingly, theconverter controller 230 may be configured as the converter controller230C shown in FIG. 30 or the converter controller 230E shown in FIG. 33.

The frequency setting circuit 290 changes the target frequency accordingto the number of the on-state current sources (the number of turned-onlight sources) from among the multiple current sources 210. Morespecifically, when the number of the on-state current sources becomessmaller than a predetermined threshold value, judgment is made that thelight load state has been detected. In this state, the frequency settingcircuit 290 sets the target frequency to a different frequency valuethat is lower than the original target frequency and does not belong toa particular band defined as an electromagnetic noise band. In a case inwhich, in the normal state, the target frequency is set to a frequencyvalue of 300 kHz to 450 kHz between the LW band and AM band, when theoperating state becomes the light load state, the target frequency maypreferably be set to a band (e.g., 100 kHz) that is lower than the LWband and that is higher than the audible band.

With an arrangement shown in FIG. 30 or 33 , the target frequency isdetermined based on the reference voltage V_(FREQ(REF)). Accordingly, ina state in which the number of the turned-on light sources is smallerthan a predetermined threshold, the frequency setting circuit 290 maypreferably reduce the reference voltage V_(FREQ(REF)).

With the embodiment 8, when the frequency is lowered in the light loadstate, such an arrangement is capable of maintaining the frequency suchthat it is outside the frequency range that causes an electromagneticnoise problem that is to be avoided.

Embodiment 9

FIG. 42 is a block diagram showing an automotive lamp 100Y according anembodiment 9. The automotive lamp 100Y further includes a dummy load 292and a dummy load control circuit 294 in addition to the configuration ofthe automotive lamp 100 shown in FIG. 25 .

The dummy load 292 is coupled to the output of the switching converter220. In the enable state, the dummy load 292 discharges the capacitor C₁of the switching converter 220 so as to lower the output voltageV_(OUT). The dummy load control circuit 294 controls the enable/disablestate of the dummy load 292 based on the number of the on-state currentsources from among the multiple current sources.

The dummy load 292 includes a switch configured as a transistor arrangedbetween the output of the switching converter 220 and the ground. Aftera predetermined time τ elapses from the turning-off of the switchingtransistor M₁, the dummy load control circuit 294 asserts (sets to thehigh level, for example) the enable signal EN, so as to turn on theswitch of the dummy load 292.

FIG. 43 is an operation waveform diagram showing the operation of theautomotive lamp 100Y shown in FIG. 42 . When the operating state becomesthe light load state, the enable signal EN is asserted for each cycle,which immediately decreases the output voltage V_(OUT). Subsequently,when the output voltage V_(OUT) decreases to a voltage level thatcorresponds to the bottom limit voltage V_(BOTTOM), the control pulse S₁is set to the high level. That is to say, the upper limit of the offtime T_(OFF) of the switching transistor M₁ is limited by thepredetermined period τ. This restricts the reduction in the switchingfrequency in the light load state.

The dummy load 292 may be configured as a constant current source thatis capable of switching its state between the on state and the offstate. Also, the dummy load 292 may be configured as a combination ofswitches and resistors.

Embodiment 10

Description will be made with reference to FIG. 25 . Typically, there isa tradeoff relation between the on resistance and the breakdown voltageof a transistor. When overshoot occurs in the output voltage V_(OUT) ofthe switching converter, this raise the voltage applied to a transistorthat forms each current source 210. Accordingly, there is a need toconfigure each current source 210 using a high-breakdown-voltageelement. However, such a high-breakdown-voltage element has a large onresistance R_(ON). Accordingly, such an arrangement requires the bottomlimit voltage V_(BOTTOM) to be set to a high value. This leads toproblems of large power consumption and large heat generation.

FIG. 44 is a circuit diagram showing a lighting circuit 200Z accordingto an embodiment 10. When the driving voltage V_(OUT) exceeds apredetermined threshold value V_(TH), the lighting circuit 200Z forciblyturns off the switching transistor M₁. The lighting circuit 200Zincludes resistors R₃₁ and R₃₂, and a voltage comparator 238. Thevoltage comparator 238 compares the driving voltage V_(OUT)′ divided bythe resistors R₃₁ and R₃₂ with a threshold value V_(TH)′, so as todetect the occurrence of an overvoltage state in the driving voltageV_(OUT).

The converter controller 230P includes a pulse modulator 235, a logicgate 233, and a driver 232. The pulse modulator 235 has the sameconfiguration as those of the converter controllers 230A through 230Eshown in FIGS. 28 through 34 except for the driver 232. The pulsemodulator 235 generates the control pulse S₁′. When the output S₂ of thevoltage comparator 238 indicates the relation V_(OUT)′<V_(TH)′, thelogic gate 233 allows the control pulse S₁′ to pass through as it is.Conversely, when the output S₂ of the voltage comparator 238 indicatesthe relation V_(OUT)′>V_(TH)′, the logic gate 233 forcibly sets thelevel of the control pulse S₁′ to a level that turns off the switchingtransistor M₁. In this example, the switching transistor M₁ isconfigured as an N-channel MOSFET. When S₁ is set to the low level, theswitching transistor M₁ is set to the off state. When V_(OUT)′>V_(TH)′,the output S₂ of the voltage comparator 238 is set to the low level. Thelogic gate 233 is configured as an AND gate.

With the present embodiment, the current source 210 is configured usinga transistor having a low on resistance, thereby allowing powerconsumption to be reduced. As a tradeoff, such an arrangement involvessuch a transistor having a low breakdown voltage. However, when anovershoot occurs in the output voltage V_(OUT) of the switchingconverter, the switching transistor M₁ is immediately suspended. Such anarrangement is capable of preventing an overvoltage from being appliedto the transistor of the current source (e.g., the transistor M₂ shownin FIGS. 36A and 36B, the output-side transistor of the current mirrorcircuit 216 shown in FIG. 36C).

Description has been made in the embodiments regarding an arrangement inwhich the current source 210 is configured as a sink circuit, and iscoupled to the cathode of the corresponding semiconductor light source102. However, the present invention is not restricted to such anarrangement. FIG. 45 is a circuit diagram showing an automotive lamp 100according to a modification. In this modification, the cathodes of thesemiconductor light sources 102 are coupled so as to form a commoncathode. Furthermore, each current source 210 configured as a sourcecircuit is coupled to the anode side of the corresponding semiconductorlight source 102. Each current source 210 may be configured bygeometrically reversing the configuration shown in FIG. 25 (or FIG. 35).

Description has been made regarding the present invention with referenceto the embodiments using specific terms. However, the above-describedembodiments show only the mechanisms and applications of the presentinvention for exemplary purposes only, and are by no means intended tobe interpreted restrictively. Rather, various modifications and variouschanges in the layout can be made without departing from the spirit andscope of the present invention defined in appended claims.

The invention claimed is:
 1. A current driver circuit structured todrive a plurality of semiconductor light sources, the current drivercircuit comprising: a plurality of current sources each structured toallow an on/off state thereof to be controlled independently accordingto a PWM signal, and to each be coupled to a corresponding semiconductorlight source in series; an interface circuit structured to receive, froman external processor at a first time interval T1, a plurality ofcontrol data that indicates an on/off duty cycle for the plurality ofcurrent sources; and a dimming pulse generator structured to generate aplurality of PWM signals for the plurality of current sources, and togradually change, at a second time interval T2 that is shorter than thefirst time interval T1, a duty cycle of each of the plurality of PWMsignals from a value indicated by the corresponding control data beforeupdating to a value indicated by the corresponding control data afterupdating, wherein, the dimming pulse generator is switchable between afirst mode and a second mode according to setting data, and wherein,with T1/T2=a, and with the difference in the value between the controldata after updating and the control data before updating as ΔX, (i) inthe first mode, the dimming pulse generator changes the duty cycle atthe second time interval T2 in a stepwise manner in steps of ΔY=ΔX/a,and (ii) in the second mode, the dimming pulse generator immediatelychanges the duty cycle from a value indicated by the correspondingcontrol data before updating to a value indicated by the correspondingcontrol data after updating.
 2. The current driver circuit according toclaim 1, wherein each of the plurality of current sources comprises: aseries transistor and a sensing resistor arranged in series with acorresponding semiconductor light source; an error amplifier structuredto adjust a voltage of a control electrode of the series transistorbased on a voltage drop that occurs across the sensing resistor; and aPWM switch arranged between a gate and a source of the seriestransistor.
 3. The current driver circuit according to claim 1, whereinthe first mode and the second mode are switchable for each of theplurality of current sources independently.
 4. The current drivercircuit according to claim 1, wherein the setting data is appended tothe control data.